Richtige Fernseher haben Röhren!

Richtige Fernseher haben Röhren!

In Brief: On this site you will find pictures and information about some of the electronic, electrical and electrotechnical technology relics that the Frank Sharp Private museum has accumulated over the years .

Premise: There are lots of vintage electrical and electronic items that have not survived well or even completely disappeared and forgotten.

Or are not being collected nowadays in proportion to their significance or prevalence in their heyday, this is bad and the main part of the death land. The heavy, ugly sarcophagus; models with few endearing qualities, devices that have some over-riding disadvantage to ownership such as heavy weight,toxicity or inflated value when dismantled, tend to be under-represented by all but the most comprehensive collections and museums. They get relegated to the bottom of the wants list, derided as 'more trouble than they are worth', or just forgotten entirely. As a result, I started to notice gaps in the current representation of the history of electronic and electrical technology to the interested member of the public.


Following this idea around a bit, convinced me that a collection of the peculiar alone could not hope to survive on its own merits, but a museum that gave equal display space to the popular and the unpopular, would bring things to the attention of the average person that he has previously passed by or been shielded from. It's a matter of culture. From this, the Obsolete Technology Tellye Web Museum concept developed and all my other things too. It's an open platform for all electrical Electronic TV technology to have its few, but NOT last, moments of fame in a working, hand-on environment. We'll never own Colossus or Faraday's first transformer, but I can show things that you can't see at the Science Museum, and let you play with things that the Smithsonian can't allow people to touch, because my remit is different.

There was a society once that was the polar opposite of our disposable, junk society. A whole nation was built on the idea of placing quality before quantity in all things. The goal was not “more and newer,” but “better and higher" .This attitude was reflected not only in the manufacturing of material goods, but also in the realms of art and architecture, as well as in the social fabric of everyday life. The goal was for each new cohort of children to stand on a higher level than the preceding cohort: they were to be healthier, stronger, more intelligent, and more vibrant in every way.

The society that prioritized human, social and material quality is a Winner. Truly, it is the high point of all Western civilization. Consequently, its defeat meant the defeat of civilization itself.

Today, the West is headed for the abyss. For the ultimate fate of our disposable society is for that society itself to be disposed of. And this will happen sooner, rather than later.

OLD, but ORIGINAL, Well made, Funny, Not remotely controlled............. and not Made in CHINA.

How to use the site:

- If you landed here via any Search Engine, you will get what you searched for and you can search more using the search this blog feature provided by Google. You can visit more posts scrolling the left blog archive of all posts of the month/year,
or you can click on the main photo-page to start from the main page. Doing so it starts from the most recent post to the older post simple clicking on the Older Post button on the bottom of each page after reading , post after post.

You can even visit all posts, time to time, when reaching the bottom end of each page and click on the Older Post button.

- If you arrived here at the main page via bookmark you can visit all the site scrolling the left blog archive of all posts of the month/year pointing were you want , or more simple You can even visit all blog posts, from newer to older, clicking at the end of each bottom page on the Older Post button.
So you can see all the blog/site content surfing all pages in it.

- The search this blog feature provided by Google is a real search engine. If you're pointing particular things it will search IT for you; or you can place a brand name in the search query at your choice and visit all results page by page. It's useful since the content of the site is very large.

Note that if you don't find what you searched for, try it after a period of time; the site is a never ending job !

Every CRT Television saved let revive knowledge, thoughts, moments of the past life which will never return again.........

Many contemporary "televisions" (more correctly named as displays) would not have this level of staying power, many would ware out or require major services within just five years or less and of course, there is that perennial bug bear of planned obsolescence where components are deliberately designed to fail and, or manufactured with limited edition specificities..... and without considering........picture......sound........quality........

..............The bitterness of poor quality is remembered long after the sweetness of todays funny gadgets low price has faded from memory........ . . . . . .....
Don't forget the past, the end of the world is upon us! Pretty soon it will all turn to dust!

Have big FUN ! !
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©2010, 2011, 2012, 2013, 2014 Frank Sharp - You do not have permission to copy photos and words from this blog, and any content may be never used it for auctions or commercial purposes, however feel free to post anything you see here with a courtesy link back, btw a link to the original post here , is mandatory.
All sets and apparates appearing here are property of
Engineer Frank Sharp. NOTHING HERE IS FOR SALE !

Sunday, December 1, 2013

ULTRAVOX (SEMAR) P12 YEAR 1985.




The ULTRAVOX (SEMAR)  P12 is a portable  12 inches (31cm) B/W television with 16 programs manual rotatable program changer and VST electronic tuning semi automatic search.

The tunign system consist in a preset tuner adapted for selecting a desired one out of a plurality of preset channels, comprising: a memory for storing digital data concerning a plurality of channels to be preset, push-buttons/rotatable knob for addressing the memory for reading the digital data of a desired channel, a digital/analog converter for converting the read digital data into an analog signal, a manually operable variable voltage generator to obtain search mode AFC assisted, a write-in/channel select mode selector to store broadcasted transmission in desired program postion by rotatable program positions knob, a switch circuit responsive to the mode selector switchable between the digital/analog converter and the variable voltage generator, a voltage controlled oscillator responsive to the output of the switch circuit, a tuner employing the voltage controlled oscillator as a local oscillator, a comparator for comparing the outputs of the digital/analog converter and the variable voltage generator,

It has a Transistorized horizontal deflection circuits  made up of a horizontal switching or output transistor, a diode, one or more capacitors and a deflection winding. The output transistor, operating as a switch, is driven by a horizontal rate square wave signal and conducts during a portion of the horizontal trace interval. A diode, connected in parallel with the transistor, conducts during the remainder of the trace interval. A retrace capacitor and the deflection yoke winding are coupled in parallel across the transistor-diode combination. Energy is transferred into and out of the deflection winding via the diode and output transistor during the trace interval and via the retrace capacitor during the retrace interval.
In some television receivers, the collector of the horizontal output transistor is coupled to the B+ power supply through the primary windings of the high voltage transformer.

Was one of the last ULTRAVOX Tv sets and last B/W of that type.


ULTRAVOX S.r.l. RADIO TELEVISIONE — Milano, 

Founded in Milan in 1948 with the denomination Ultravox S.r.l of the brothers Sergio and Cino Stanghi, it was dedicated initially to the production of radiophonic apparatus.

 In 1959 the production was moved in a factory to Caronno Pertusella, in province of Varese, where some years later, there was transferred also the seat of the society, which changed in Ultravox S.p.A.

Later  it was specialised in the television sets production, sector in which the mark Ultravox became celebrated and conquered spaces in the market, with the carrying out of a special portable television set completely to transistor with tuner I expel incorporated FM, the model Hummingbird with protection from 6 thumbs of 1969, planned at the designer's Giovanni's Offredi.

In general the models Ultravox were distinguished compared to others, for the very original and futurist design, like the record player Cockchafer of 1970, created always from Offredi. Particular was the models of television sets built with incorporated record player.

In the eighties, the Lombard company, it had interested  a crisis in the market of the electronics that put it in financial difficulties, and therefore in 1982 it was admitted in the list of the societies in field REL, public society for the curing of the enterprises of the electronic sector.

The REL entered into the capital of the Ultravox acquiring 27 % it, since in 1981, the company, which was counting 105 employees, recorded losses in the budget accumulated 1970, piling up to 926 million lire. The accounts improved in the following years since the turnover passed 2,4 milliards from 1982 to 36 in 1987, year in which after years of losses it reached a profit of budget.

In 1988 together with the REL there acquired the control of Emerson of Florence, company television sets producer. They were noticed the mark of the society and the factory of Siena in resort It isolates of Arbia, when the Ultravox was constituting Siena S.p. A. The Sienese factory was reactivated and there was guided the production of television sets and videotape recorders with the marks Ultravox and Emerson.

In the nineties, the company was swept away by a new crisis, which was heavier than that one that struck it on the antecedent decade.

In 1993 it was forced the factory of Caronno Pertusella to close and subsequently it went to checked administration. To go out of this situation, in the capital of the Ultravox there intervened the Bank Mountain of the Paschi of Siena, with a financing for 8,5 milliard lires.

The Sienese factory remained active only with his 200 employees, and the Ultravox was taken from the Holding company Galaxis, German financial society that was checking the Galaxis, company decoder producer for the reception of satellite TV. Initially the Tuscan society unwound the list of Italian distributor for the Galaxis, then later on it specialised in the production of the decoder. Under the new property the company grew again be in terms of turnover in which of sales, and in 1996, Ultravox became the principal decoder supplier for the television paytv Telepiù (died).

In 1998 the Sienese company changed social reason in Galaxis Produzione S.p. A., and it was producing television sets, videotape recorders and decoder for pay TV with the marks Ultravox and Galaxis.

In 2000 the Ultravox-Galaxis is forced to the closing for bankruptcy.


ULTRAVOX (SEMAR) P12 CHASSIS 2T/1 INTERNAL VIEW.









Power supply is realized with mains transformer and Linear transistorized power supply stabilizer, A DC power supply apparatus includes a rectifier circuit which rectifies an input commercial AC voltage. The rectifier output voltage is smoothed in a smoothing capacitor. Voltage stabilization is provided in the stabilizing circuits by the use of Zener diode circuits to provide biasing to control the collector-emitter paths of respective transistors.A linear regulator circuit according to an embodiment of the present invention has an input node receiving an unregulated voltage and an output node providing a regulated voltage. The linear regulator circuit includes a voltage regulator, a bias circuit, and a current control device.

In one embodiment, the current control device is implemented as an NPN bipolar junction transistor (BJT) having a collector electrode forming the input node of the linear regulator circuit, an emitter electrode coupled to the input of the voltage regulator, and a base electrode coupled to the second terminal of the bias circuit. A first capacitor may be coupled between the input and reference terminals of the voltage regulator and a second capacitor may be coupled between the output and reference terminals of the voltage regulator. The voltage regulator may be implemented as known to those skilled in the art, such as an LDO or non-LDO 3-terminal regulator or the like.
The bias circuit may include a bias device and a current source. The bias device has a first terminal coupled to the output terminal of the voltage regulator and a second terminal coupled to the control electrode of the current control device. The current source has an input coupled to the first current electrode of the current control device and an output coupled to the second terminal of the bias device. A capacitor may be coupled between the first and second terminals of the bias device.
In the bias device and current source embodiment, the bias device may be implemented as a Zener diode, one or more diodes coupled in series, at least one light emitting diode, or any other bias device which develops sufficient voltage while receiving current from the current source. The current source may be implemented with a PNP BJT having its collector electrode coupled to the second terminal of the bias device, at least one first resistor having a first end coupled to the emitter electrode of the PNP BJT and a second end, a Zener diode and a second resistor. The Zener diode has an anode coupled to the base electrode of the PNP BJT and a cathode coupled to the second end of the first resistor. The second resistor has a first end coupled to the anode of the Zener diode and a second end coupled to the reference terminal of the voltage regulator. A second Zener diode may be included having an anode coupled to the cathode of the first Zener diode and a cathode coupled to the first current electrode of the current control device.
A circuit is disclosed for improving operation of a linear regulator, having an input terminal, an output terminal, and a reference terminal. The circuit includes an input node, a transistor, a bias circuit, and first and second capacitors. The transistor has a first current electrode coupled to the input node, a second current electrode for coupling to the input terminal of the linear regulator, and a control electrode. The bias circuit has a first terminal for coupling to the output terminal of the linear regulator and a second terminal coupled to the control electrode of the transistor. The first capacitor is for coupling between the input and reference terminals of the linear regulator, and the second capacitor is for coupling between the output and reference terminals of the linear regulator. The bias circuit develops a voltage sufficient to drive the control terminal of the transistor and to operate the linear regulator. The bias circuit may be a battery, a bias device and a current source, a floating power supply, a charge pump, or any combination thereof. The transistor may be implemented as a BJT or FET or any other suitable current controlled device.



Power Supply: The examples chosen are taken from manufacturers' circuit diagrams and are usually simplified to emphasise the fundamental nature of the circuit. For each example the particular transistor properties that are exploited to achieve the desired performance are made clear. As a rough and ready classification the circuits are arranged in order of frequency: this part is devoted to circuits used at zero frequency, field frequency and audio frequencies. Series Regulator Circuit Portable television receivers are designed to operate from batteries (usually 12V car batteries) and from the a.c. mains. The receiver usually has an 11V supply line, and circuitry is required to ensure that the supply line is at this voltage whether the power source is a battery or the mains. The supply line also needs to have good regulation, i.e. a low output resistance, to ensure that the voltage remains constant in spite of variations in the mean current taken by some of the stages in the receiver. Fig. 1 shows a typical circuit of the power -supply arrangements. The mains transformer and bridge rectifier are designed to deliver about 16V. The battery can be assumed to give just over 12V. Both feed the regulator circuit Trl, Tr2, Tr3, which gives an 11V output and can be regarded as a three -stage direct -coupled amplifier. The first stage Tr 1 is required to give an output current proportional to the difference between two voltages, one being a constant voltage derived from the voltage reference diode D I (which is biased via R3 from the stabilised supply). The second voltage is obtained from a preset potential divider connected across the output of the unit, and is therefore a sample of the output voltage. In effect therefore Tr 1 compares the output voltage of the unit with a fixed voltage and gives an output current proportional to the difference between them. Clearly a field-effect transistor could do this, but the low input resistance of a bipolar transistor is no disadvantage and it can give a current output many times that of a field-effect transistor and is generally preferred therefore. The output current of the first stage is amplified by the two subsequent stages and then becomes the output current of the unit. Clearly therefore Tr2 and Tr3 should be current amplifiers and they normally take the form of emitter followers or common emitter stages (which have the same current gain). By adjusting the preset control we can alter the fraction of the output voltage' applied to the first stage and can thus set the output voltage of the unit at any desired value within a certain range. By making assumptions about the current gain of the transistors we can calculate the degree of regulation obtainable. For example, suppose the gain of Tr2 and Tr3 in cascade is 1,000, and that the current output demanded from the unit changes by 0.1A (for example due to the disconnection of part of the load). The corresponding change in Tr l's collector current is 0.1mA and, if the standing collector current of Tr 1 is 1mA, then its mutual conductance is approximately 4OmA/V and the base voltage must change by 2.5mV to bring about the required change in collector current. If the preset potential divider feeds one half of the output voltage to Tr l's base, then the change in output voltage must be 5mV. Thus an 0.1A change in output current brings about only 5mV change in output voltage: this represents an output resistance of only 0.0552.

GENERAL BASIC TRANSISTOR LINE OUTPUT STAGE OPERATION:

The basic essentials of a transistor line output stage are shown in Fig. 1(a). They comprise: a line output transformer which provides the d.c. feed to the line output transistor and serves mainly to generate the high -voltage pulse from which the e.h.t. is derived, and also in practice other supplies for various sections of the receiver; the line output transistor and its parallel efficiency diode which form a bidirectional switch; a tuning capacitor which resonates with the line output transformer primary winding and the scan coils to determine the flyback time; and the scan coils, with a series capacitor which provides a d.c. block and also serves to provide slight integration of the deflection current to compensate for the scan distortion that would otherwise be present due to the use of flat screen, wide deflection angle c.r.t.s. This basic circuit is widely used in small -screen portable receivers with little elaboration - some use a pnp output transistor however, with its collector connected to chassis.

Circuit Variations:
Variations to the basic circuit commonly found include: transposition of the scan coils and the correction capacitor; connection of the line output transformer primary winding and its e.h.t. overwinding in series; connection of the deflection components to a tap on the transformer to obtain correct matching of the components and conditions in the stage; use of a boost diode which operates in identical manner to the arrangement used in valve line output stages, thereby increasing the effective supply to the stage; omission of the efficiency diode where the stage is operated from an h.t. line, the collector -base junction of the line output transistor then providing the efficiency diode action without, in doing so, producing scan distortion; addition of inductors to provide linearity and width adjustment; use of a pair of series -connected line output transistors in some large -screen colour chassis; and in colour sets the addition of line convergence circuitry which is normally connected in series between the line scan coils and chassis. These variations on the basic circuit do not alter the basic mode of operation however.

Resonance
The most important fact to appreciate about the circuit is that when the transistor and diode are cut off during the flyback period - when the beam is being rapidly returned from the right-hand side of the screen to the left-hand side the tuning capacitor together with the scan coils and the primary winding of the line output transformer form a parallel resonant circuit: the equivalent circuit is shown in Fig. 1(b). The line output transformer primary winding and the tuning capacitor as drawn in Fig. 1(a) may look like a series tuned circuit, but from the signal point of view the end of the transformer primary winding connected to the power supply is earthy, giving the equivalent arrangement shown in Fig. 1(b).

The Flyback Period:
Since the operation of the circuit depends mainly upon what happens during the line flyback period, the simplest point at which to break into the scanning cycle is at the end of the forward scan, i.e. with the beam deflected to the right-hand side of the screen, see Fig. 2. At this point the line output transistor is suddenly switched off by the squarewave drive applied to its base. Prior to this action a linearly increasing current has been flowing in the line output transformer primary winding and the scan coils, and as a result magnetic fields have been built up around these components. When the transistor is switched off these fields collapse, maintaining a flow of current which rapidly decays to zero and returns the beam to the centre of the screen. This flow of current charges the tuning capacitor, and the voltage at A rises to a high positive value - of the order of 1- 2k V in large -screen sets, 200V in the case of mains/battery portable sets. The energy in the circuit is now stored in the tuning capacitor which next discharges, reversing the flow of current in the circuit with the result that the beam is rapidly deflected to the left-hand side of the screen - see Fig. 3. When the tuning capacitor has discharged, the voltage at A has fallen to zero and the circuit energy is once more stored in the form of magnetic fields around the inductive components. One half -cycle of oscillation has occurred, and the flyback is complete.

Energy Recovery:
First Part of Forward Scan The circuit then tries to continue the cycle of oscillation, i.e. the magnetic fields again collapse, maintaining a current flow which this time would charge the tuning capacitor negatively (upper plate). When the voltage at A reaches about -0.6V however the efficiency diode becomes forward biased and switches on. This damps the circuit, preventing further oscillation, but the magnetic fields continue to collapse and in doing so produce a linearly decaying current flow which provides the first part of the forward scan, the beam returning towards the centre of the screen - see Fig. 4. The diode shorts out the tuning capacitor but the scan correction capacitor charges during this period, its right-hand plate becoming positive with respect to its left-hand plate, i.e. point A. Completion of Forward Scan When the current falls to zero, the diode will switch off. Shortly before this state of affairs is reached however the transistor is switched on. In practice this is usually about a third of the way through the scan. The squarewave applied to its base drives it rapidly to saturation, clamping the voltage at point A at a small positive value - the collector emitter saturation voltage of the transistor. Current now flows via the transistor and the primary winding of the line output transformer, the scan correction capacitor discharges, and the resultant flow of current in the line scan coils drives the beam to the right-hand side of the screen see Fig. 5.

Efficiency:
The transistor is then cut off again, to give the flyback, and the cycle of events recurs. The efficiency of the circuit is high since there is negligible resistance present. Energy is fed into the circuit in the form of the magnetic fields that build up when the output transistor is switched on. This action connects the line output transformer primary winding across the supply, and as a result a linearly increasing current flows through it. Since the width is
dependent on the supply voltage, this must be stabilised.

Harmonic Tuning:
There is another oscillatory action in the circuit during the flyback period. The considerable leakage inductance between the primary and the e.h.t. windings of the line output transformer, and the appreciable self -capacitance present, form a tuned circuit which is shocked into oscillation by the flyback pulse. Unless this oscillation is controlled, it will continue into and modulate the scan. The technique used to overcome this effect is to tune the leakage inductance and the associated capacitance to an odd harmonic of the line flyback oscillation frequency. By doing this the oscillatory actions present at the beginning of the scan cancel. Either third or fifth harmonic tuning is used. Third harmonic tuning also has the effect of increasing the amplitude of the e.h.t. pulse, and is generally used where a half -wave e.h.t. rectifier is employed. Fifth harmonic tuning results in a flat-topped e.h.t. pulse, giving improved e.h.t. regulation, and is generally used where an e.h.t. tripler is employed to produce the e.h.t. The tuning is mainly built into the line output transformer, though an external variable inductance is commonly found in colour chassis so that the tuning can be adjusted. With a following post I will go into the subject of modern TV line timebases in greater detail with other models and technology shown here at  Obsolete Technology Tellye !


PHILIPS TDA2549 I.F. amplifier and demodulator for multistandard TV receivers:

GENERAL DESCRIPTION
The TDA2549 is a complete i.f. circuit with a.f.c., a.g.c., demodulation and video preamplification facilities for
multistandard television receivers. It is capable of handling positively and negatively modulated video signals in both
colour and black/white receivers.
Features
• Gain-controlled wide-band amplifier providing complete i.f. gain
• Synchronous demodulator for positive and negative modulation
• Video preamplifier with noise protection for negative modulation
• Auxiliary video input and output (75 Ω)
• Video switch to select between auxiliary video input signal and demodulated video signal
• A.F.C. circuit with on/off switch and inverter switch
• A.G.C. circuit for positive modulation (mean level) and negative modulation (noise gate)
• A.G.C. output for controlling MOSFET tuners.

THOMSON TDA3190 COMPLETE TV SOUND CHANNEL:

The TDA3190 is a monolithicintegratedcircuit in a
16-lead dual in-line plastic package.It performsall
the functionsneededfor the TV soundchannel :
.IF LIMITER AMPLIFIER
.ACTIVE LOW-PASSFILTER
.FM DETECTOR
.DC VOLUMECONTROL
.AF PREAMPLIFIER
.AF OUTPUT STAGE
DESCRIPTION
The TDA3190 can give an output power of 4.2 W
(d = 10 %) into a 16 Ω load at VS = 24 V, or 1.5 W
(d = 10 %) into an 8 Ω load at VS = 12 V. This
performance,togetherwiththe FM-IF sectionchar-
acteristicsof high sensitivity, highAM rejection and
low distortion, enables the device to be used in
almost every type of televisionreceivers.
The device has no irradiation problems, hence no
externalscreening is needed.
The TDA3190 is a pin to pin replacement of
TDA1190Z.
The electrical characteristics of the TDA3190 re-
mainalmost constantover the frequencyrange4.5
to 6 MHz, therefore it can be used in all television
standards (FM mod.). The TDA3190 has a high
inputimpedance,soitcanwork withaceramicfilter
or with a tuned circuit that provide the necessary
input selectivity.
The value of the resistors connected to pin 9,
determinethe AC gain of the audio frequencyam-
plifier. This enablesthe desiredgainto be selected
in relation to the frequency deviation at which the
output stage of the AF amplifier, must enter into
clipping.
CapacitorC8, connectedbetween pins10 and11,
determinesthe uppercutofffrequencyof the audio
bandwidth.Toincreasethebandwidththe valuesof
C8 and C7 must be reduced, keeping the ratio
C7/C8 as shown in the table of fig. 16.
The capacitor connected between pin 16 and
ground, togetherwith the internal resistor of 10 KΩ
forms the de-emphasis network. The Boucherot
cell eliminates the high frequency oscillations
causedbytheinductiveloadandthewiresconnect-
ing theloudspeaker.


ULTRAVOX (SEMAR)  P12  CHASSIS 2T/1 Preset tuner:

 A preset tuner adapted for selecting a desired one out of a plurality of preset channels, comprising: a memory for storing digital data concerning a plurality of channels to be preset, push-buttons/rotatable knob for addressing the memory for reading the digital data of a desired channel, a digital/analog converter for converting the read digital data into an analog signal, a manually operable variable voltage generator to obtain search mode AFC assisted, a write-in/channel select mode selector, a switch circuit responsive to the mode selector switchable between the digital/analog converter and the variable voltage generator, a voltage controlled oscillator responsive to the output of the switch circuit, a tuner employing the voltage controlled oscillator as a local oscillator, a comparator for comparing the outputs of the digital/analog converter and the variable voltage generator, a counter to be reset responsive to the push-button search mode AFC  and to be enabled responsive to the write-in mode output of the mode selector to make a counting operation as a function of the output of the comparator, the output of the counter being loaded as the digital data concerning a channel in the memory, as addressed, a reference oscillator, a frequency comparator for comparing the output of the reference oscillator and the intermediate frequency output of the tuner, a filter for filtering the frequency comparator output to provide a correction control signal to the digital/analog converter, the digital/analog converter being adapted to be corrected for deviation of the intermediate frequency of the tuner as a function of the correction control signal.

 A typical conventional preset tuner such as employed in tv receivers and the like comprises a mechanical preset scheme. For example, such a preset tuner employing variable inductance devices as a tuning element is adapted to preset a plurality of channels by varying the inductance value of the variable inductance devices in association with the manual operation of a tuning knob. Another example of such a tuner using a variable capacitance device such as a variable capacitance diode as a tuning element employs variable resistors adjustable in association with the manual operation of a tuning knob for the purpose of a presetting operation.

 Such preset tuners as described above as employing a variable inductance device, a variable capacitance device and the like require provision of the same number of variable inductance devices, variable resistors and the like as that of presetting channel selection switches, which makes the tuner large in size, with the result that such tuner is disadvantageous particularly in case where such tuner is employed in an indash type portable tv receiver, where the tuner is provided in a limited space.


Accordingly, a principal object of the present invention is to provide an improved preset tuner, wherein the data concerning the local oscillation frequencies corresponding to the respective preset channels is preloaded in a memory in a digital representation format.
Another object of the present invention is to provide an improved preset tuner, which is adapted for implementation by large scale integration integrated circuits.
A further object of the present invention is to provide an improved preset tuner, which is adapted for implementation in an electronic structure rather than mechanical structure

ULTRAVOX (SEMAR) P12 CHASSIS 2T/1 CRT TUBE A31-510W.


TELEVISION TUBE A31-510W

QUICK REFERENCE DATA

31cm (12in) rectangular direct viewing television tube. A separate safety
screen is not required. Especially for use in portable receivers with push-
through presentation.

A special feature of this tube is its short warm—up time.

Deflection angle 110 deg
Final accelerator voltage max. 15 kV
Neck diameter 20 mm —u-

Light transmission 50 % E

Maximum overall length 233 mm

A legible picture appears within 5 seconds (typ. )

This data should be read in conjunction with
GENERAL OPERATIONAL RECOMMENDATIONS — TELEVISION PICTURE TUBES

HEATER
Vh 11  V
Ah 140 mA
Cathode warm —up time (typ. ) 5 s

OPERATING CONDITIONS

Va2, a4 12 kV
VG3 (focus electrode) control range 0 to 350 V
V  A1 250 V
Vg for visual extinction of focused raster  -35 to -69 V
*Vk for visual extinction of focused raster 32 to 58 V

*For cathode modulation, all voltages are measured with respect to grid.

SCREEN
Metal backedFluorescent colour White
Light transmission (approx. ) 50 %











Focusing Electrostatic

DEFLECTION Magnetic
Diagonal deflection ‘angle  110 deg
Horizontal deflection angle 99 deg
Vertical deflection angle 80 deg

The deflection eons should designed so that their internal contour is in accordance
with the reference line gauge shown on page 4.



CAPACITANCES:
cg all 7. 0 pF
ck all 3. 0 pF
ca2 a4 m 450 to 900 pF
a2,a4 —B 150 PF

EXTERNAL CONDUCTIVE COATING
This tube has external conductive coating, M, which must be connected to chassis.
and the 'capacitance ‘of this ‘coating to the final anode is used to provide smoothing
for the eht supply.
 The electrical connection to this coating must be made within
the area specified on the tube outline drawing;

RASTER CENTRING

See notes under this heading in ' General Operational Recommendations - Television
Picture Tubes' .

Centring magnet field intensity 0 to 800 A/m
Maximum distance of centre of centering field from reference line 47 mm

Adjustment of the centring magnet should not be such that '3. general reduction in
brightness of the raster occurs.

REFERENCE LINE GAUGE see page 4
MOUNTING POSITION Any

The tube socket should not be rigidily mounted but should have flexible leads and be
allowed to move freely.

This ‘tube is fitted with a pin protector in "order to avoid ‘damage to ‘the glass base
due to bending of the base pins whilst handling the tube.

it is advisable to keep this pin protector on the base until it can be replaced by the
socket after the installation of the tube in any equipment.

TELEVISION TUBE A31-410W

RATINGS (DESIGN MAXIMUM SYSTEM)

Va2 a4 max. (at i=0 a2 a4) (see note 1) 15 kV
Vva2 a4,“ min, 8.5kv
va3 max. 500 V
-Va3 max. 50 V
Va1 max. 350 V
va1 min. 200 V
~vg(pk) max. (see note 2) 350 V
~Vg max. (see note 3) ~ 100 V



Adequate precautions should be taken to ensure that the receiver is protected
from damage which may be caused by a possible high voltage flashover within
the tube. '

Maximum pulse duration 22% of one cycle with a maximum of 1. 5ms

The d. c. value of bias must not be such as to allow the grid to become positive
with respect to the cathode, except during the period immediately after switching
the receiver on or off when it may be allowed to rise to +2. OV. It is advisable to
limit the positive excursion of the video signal to +5V(pk) max. This may be
achieved automatically by the series connection of a 10kohm resistor.

. The metal hand must be earthed by means of the tag provided.

The mounting lugs will not necessarily be in electrical contact with the metal
band.

Weight tube alone (approx. ) 2. 8 kg




Saturday, November 30, 2013

NORDMENDE (THOMSON) CONTURA 63 (ICC9) YEAR 1993.






 The NORDMENDE (THOMSON) CONTURA 63  is a 25 inches color television with planar square screen, black matrix.

The set features mutistandard feature, stereo hifi sound, teletext, CTI, LTI, SVHS, 2 AV SCART sockets, external loudpseakers jacks, headphone jack, osd...............

The Teletext is a television-based communication technique in which a given horizontal video line is utilized for broadcasting textual and graphical information encoded in a digital binary representation. Such horizontal video line signal that contains teletext data is referred to herein as a Data-line. It is assumed herein, for explanation purposes, that teletext is sent by the broadcaster only during the vertical blanking interval (VBI), when no other picture information is sent. The organization of the binary information in the broadcast signal is determined by the standard employed by the broadcaster. By way of an example only, references are made herein to a teletext based on a standard referred to by the British Broadcasting Corporation (BBC) as CEEFAX.

Each Data-line carries data synchronizing and address information and the codes for a Row of 40 characters. The synchronizing information includes a clock run-in sequence followed by an 8-bit framing code sequence. Each Data-line contains a 3 bit code referred to as the Magazine number. A teletext Page includes 24 Rows of 40 characters, including a special top Row called the Page-Header. Each ROW is contained in a corresponding Data-line. A user selected Page is intended to be displayed in place of, or added to a corresponding television picture frame. A Magazine is defined to include Pages having Data-lines containing a corresponding Magazine number. The transmission of a selected Page begins with, and includes its Page Header and ends with and excludes the next Page Header of the selected Magazine number. All intermediate Data lines carrying the selected Magazine number relate to the selected Page.
A SCART Connector (which stands for Syndicat des Constructeurs d'Appareils Radiorécepteurs et Téléviseurs) is a standard for connecting audio-visual equipment together. The official standard for SCART is CENELEC document number EN 50049-1. SCART is also known as Péritel (especially in France) and Euroconnector but the name SCART will be used exclusively herein. The standard defines a 21-pin connector (herein after a SCART connector) for carrying analog television signals. Various pieces of equipment may be connected by cables having a plug fitting the SCART connectors. Television apparatuses commonly include one or more SCART connectors.
Although a SCART connector is bidirectional, the present invention is concerned with the use of a SCART connector as an input connector for receiving signals into a television apparatus. A SCART connector can receive input television signals either in an RGB format in which the red, green and blue signals are received on Pins 15, 11 and 7, respectively, or alternatively in an S-Video format in which the luminance (Y) and chroma (C) signals are received on Pins 20 and 15. As a result of the common usage of Pin 15 in accordance with the SCART standard, a SCART connector cannot receive input television signals in an RGB format and in an S-Video format at the same time.
Consequently many commercially available television apparatuses include a separate SCART connectors each dedicated to receive input television signals in one of an RGB format and an S-Video format. This limits the functionality of the SCART connectors. In practical terms, the number of SCART connectors which can be provided on a television apparatus is limited by cost and space considerations. However, different users wish the input a wide range of different combinations of formats of television signals, depending on the equipment they personally own and use. However, the provision of SCART connectors dedicated to input television signals in one of an RGB format and an S-Video format limits the overall connectivity of the television apparatus. Furthermore, for many users the different RGB format and S-Video format are confusing. Some users may not understand or may mistake the format of a television signal being supplied on a given cable from a given piece of equipment. This can result in the supply of input television signals of an inappropriate format for the SCART connector concerned.
This kind of connector is todays obsoleted !

The set was first tv set from THOMSON featuring the sophisticated chassis ICC9 wich was pretty unique and special by design and technology and  complications THOMSON STYLE.

And was first set introducing the picture signal improvement. A method of picture signal improvement in accordance with a first aspect of the invention, an input picture signal is filtered to obtain a high-frequency component and a low-frequency component, a standard deviation signal is calculated from the input picture signal, the high-frequency component is gain controlled in dependence upon the standard deviation signal to obtain a gain-controlled high-frequency component, and the gain-controlled high-frequency component and the low-frequency component are combined to obtain an output signal.
Prior means for improving the edge transitions of video signals include the addition of an enhancement or peaking signal to the original video signal. These enhancement or peaking signals are normally generated through such techniques as taking the first or second derivative of the original video signal, and adding the derivative signal, in an appropriate amount and polarity, to the original video signal. Another technique is to use a transversal filter, such as a tapped delay line; wherein, input, output and tap signals are suitably weighted, signed and combined to produce a "peaked" output signal. Still another technique is use a delay line that is unterminated on its output but terminated on its input. By subtracting the output signal from the input signal, a "peaking" signal is produced that is added in an appropriate amount and polarity to the output signal.
 In order for these methods to achieve significant improvement in the resulting signals rise and fall times of edge transitions, a relatively large amount of enhancement or "peaking" signal must be added to the original video signal. However, this also produces excessive preshoots and overshoots at the edge transitions. In a television system, the white going preshoots and overshoots can cause spot blooming on the CRT display. It is an object of the present invention to furnish apparatus for sharpening the edges, i.e. decreasing the transition times from a dark to a light level and vice versa in a received video signal in which such changes take place at varying rates, without adding preshoots or overshoots to the resultant signal.

It is a further object of the present invention to carry out this objective with a minimum of additional circuitry in a reliable fashion.

According to the present invention, the first and second derivatives of the incoming signal are generated, and the so generated derivatives are multiplied to each other to generate a control signal.

The incoming signal is also subjected to a first and second time delay. A switching circuit operative under control of the control signal normally connects the signal at the output of the first time delay to the circuit output. In response to the control signal, the circuit output is first connected to the output of the second time delay, next to the signal directly as received and finally back to the signal at the output of the first time delay.

The second time delay exceeds the first time delay and, preferably, is equal to twice the first time delay. The sharpening of the edges thus occurs because the transition is delayed until th directly received video signal has completed the transition. At this point the signal output is switched to the direct input, causing a sharp transition. After the final value of the transition has been stabilized, the signal output is switched back to the output of the first time delay where it remains until the next subsequent transition takes place.

In a preferred embodiment of the invention, the control signal is derived by multiplying the first and second derivatives of the incoming signal to each other and amplitude limiting the resulting signal.

Introduces even the CTI feature for the sharpening of colour transitions. 
In the past analog commercial TV transmission standards, the limited bandwidth of the transmitted chrominance (or chrominance difference) signals causes the received images to have perceptibly blurred colour transition edges. This is especially evident if the received image contains geometrical patterns, e.g. test-colour bars, and results in the loss of detail detectable in complex multicoloured fine patterns.

In order to improve the quality of the received images, it is necessary to provide the receiver end with circuits capable of restoring, as far as possible, the frequency components in the chrominance signals which have been filtered away by the requirements of the reduced transmission bandwidth: in this way, the temporal duration of the chrominance transition edges, and thus the spatial extent of the chrominance transitions on the TV screen, can be reduced, and the edge definition improved. Circuits of this type are called "Color Transient Improvement" ("CTI") or Chrominance Transition Enhancement circuits.

An important constraint on chrominance transition enhancement circuits is the need to ensure that the center of the chrominance transition is unaffected by the enhancement process, so that the center of the chrominance transition after the enhancement process is still aligned with the center of the associated transition in the luminance signal. Also, it is necessary to leave gradual transitions in time unaltered; preserve, and possibly enhance, fine patterns; prevent the introduction in the image of additional distortions; and ensure that the existing noise components are not accentuated.
The set has superb pictures with very precise picture contour and crisp colors thankfully to special Signal processings.

The set is last series with PLANAR CRT tube after that was abandoned definitely.

 The VIDEOCOLOR PLANAR featured has even an improvement for increased EHT to obtain more focussing.One aspect of the present invention is concerned with reducing the size of the crossover, and thus of the image thereof on the screen, compared with the known gun. In accordance with this aspect of the invention, the voltage applied to the first anode is higher than in a corresponding conventional gun and in particular is greater than the voltage applied to the focussing anode. As a result, a high electric field is formed between the grid and the first anode which tends to reduce the size of the crossover, aspect of the invention seeks to utilise this high voltage in controlling the beam size.



(To see the Internal Chassis Just click on Older Post Button on bottom page, that's simple !)




Nordmende was a manufacturer of entertainment electronics based in Bremen, Germany.
The original company, Radio H. Mende & Co, was founded in 1923 by Otto Hermann Mende (1885-1940) in Dresden. Following the destruction of the plant during the bombing raids in 1945, Martin Mende (the founder's son) created a new company in Bremen in 1947, in a former Focke-Wulf plant, under the name North German Mende Broadcast GmbH. The name was subsequently changed to Nordmende: subsequently the company became one of the prominent German manufacturers of radios, televisions, tape recorders and record players in the 1950s and 1960s.
In the 1970s, Nordmende televisions were renowned for their innovative chassis, and for the rigorous testing and quality control of their finished products. Both created high costs, however, which soon proved a competitive disadvantage when the price of colour televisions began to plunge.
In 1969, Mende's sons took over the company, and in 1977 a majority shareholding was sold to the French Thomson Brandt company and the chassis remains the original NordMende until CHASSIS F9. The following year, the family sold their remaining shares to Thomson. In the 1980s, the factories in Bremen were closed, Nordmende becoming purely a Thomson trademark (Starting from chassis F10 F11 they're all THOMSON).








In the 1990s, the name Nordmende was used with decreasing frequency, and it eventually disappeared in favour of the Thomson name. In 2005 Videocon Group acquired all cathode ray tube activities from Thomson. This led to the creation of VDC Technologies, which manufactures TV sets using the Nordmende brand under licence from Thomson.
The Nordmende brand name was relaunched in Ireland in September 2008 by the KAL Group. Although Nordmende was well known for its televisions throughout Ireland during the 1970s and 1980s, the company bought the rights to the name and launched a range of white goods including fridges, freezers, washing machines, and dishwashers, alongside a revamped range of flat-screen TVs and stereos.


     
NORDMENDE HISTORY IN GERMAN:

Die Vorkriegsgeschichte findet sich unter Mende. Nach dem Totalverlust in Dresden gründet Martin Mende (30.12.1898-1982) unter Mitwirkung von Hermann Weber am 26. August 1947 [FT5901] in Bremen-Hemelingen die Norddeutsche Mende-Rundfunk GmbH.

Die ersten Gehäuse liefert ein Tischler in Achim gegen Kompensation von fünf Gehäusen zu einem Rundfunkgerät. Der frühere Mende-Konstrukteur, Obering. Heer zeichnet wieder für die Geräte verantwortlich [FT49??].

Ab 27. Juli 1948 liefert die neue, zuerst 18 und bald 60 Personen umfassende Firma auf Grund von Währungsreform, Krediten und Zulieferverträgen die neue Radioproduktion.

Das Regime in Ostdeutschland lässt den Namen Mende nicht zu, so dass Martin Mende mit grafischen Konstruktionen im Zusammenhang mit «Nord» an seinen Vorkriegserfolg anschliesst.

Die Hallen der ehemaligen Focke-Wulf AG beim Bahnhof Seebaldsbrück dienen als Werkstätten. 1950 beschäftigt das Unternehmen 700, 1959 schon 3500 und im Zenit 6300 Personen.

1950 beginnt die Firma mit UKW-, 1953 mit Fernseh- und 1954 mit Mess- und Prüfgeräten. Gegen Ende der 50er Jahre heisst die Firma Norddeutsche Mende Rundfunk KG [RP7901].

Nachdem sich Nordmende bislang nicht mit Magnettongeräten befasst hat, bringt das Werk 1958 das erste deutsche Heim-Tonbandgerät mit drei Motoren auf den Markt. Allerdings dominieren auf diesem Sektor eindeutig andere Firmen wie AEG/Telefunken und Grundig. Von Nordmende kommen jeweils nur ein bis zwei Geräte (1960 keines) in die Kataloge. Dafür hat die Firma Erfolg mit einem anderen Neueinstieg:


1958 stellt Nordmende mit «Mambo» ihr erstes Reisegerät vor - aber nicht «das erste deutsche, serienmässig hergestellte und volltransistorisierte Koffergerät», wie man aus einer Quelle nachlesen kann. Danach wird Nordmende in Deutschland auf dem Sektor Reisegeräte besonders stark, obwohl sie keine Röhren-Koffer baute. Immerhin kosten die in «Mambo» verwendeten 8 Halbleiter dann im Einzelhandel DM 98.70, während für die vier D-Röhren der 90er-Serie - auch zum Katalogpreis - etwa DM 35.- auszugeben wären. Preis des ganzen Gerätes: DM 189.- plus zwei Flachbatterien von 4,5 V.

Bis 1969 gibt es ca. 92 Modelle der tragbaren Radios (Koffer- bzw. «Handradios», d.h. «Hand held radios»). Beispielsweise finden sich im Katalog 1961/62 [448] je 11 Tischradios und Radiomöbel sowie 8 Modelle von Reiseradios. 17 verschiedene Fernsehmodelle zeigen dagegen, wo in jener Zeit der Erfolg zu holen war.

Gemäss [FT7901] liegt Nordmende während kurzer Zeit mit der sogenannten «Tippomatik-Bedienung» sogar technisch vorne. Siehe auch Philips etc.

Auch Konzertschränke scheinen Ende der 50er bis Anfang 60er Jahre eine tragende Säule für Nordmende zu sein. Dabei verwendet die Firma immer wieder gleiche Namen wie «Cabinet», «Caruso», «Casino», «Cosima» sowie «Arabella» und «Isobella» mit wechselnden Zusatz-Nummern oder den Zusatz «Stereo», z.B. in den Jahren 1959 und 1960/61.

Im März 1967 nimmt das Werk die Produktion von Farbfernsehgeräten auf. Zum Firmenjubiläum erscheint eine Gerätereihe mit der Bezeichnung 'Goldene 20'. 1969 übernehmen die Mende-Söhne Karl und Hermann die Geschäftsführung.

1977 führt der verschärfte Wettbewerb zum Verkauf der Mehrheit an den französischen Konzern Thomson-Brandt; die Familie Mende zieht sich anschliessend ganz aus dem Unternehmen zurück. Martin Mende stirbt 1982.


In 1879 Elihu Thomson and Edwin Houston formed the Thomson-Houston Electric Company in the United States.
On April 15, 1892 Thomson-Houston and the Edison General Electric Company merged to form General Electric (GE). Also in 1892 the company formed a French subsidiary, Thomson Houston International.
In 1893 Compagnie Française Thomson-Houston (CFTH) was set up as a partner to GE. It is from this company that the modern Thomson companies would evolve.
In 1966 CFTH merged with Hotchkiss-Brandt to form Thomson-Houston-Hotchkiss-Brandt (soon renamed Thomson-Brandt). In 1968 the electronics business of Thomson-Brandt merged with Compagnie Générale de Télégraphie Sans Fil (CSF) to form Thomson-CSF. Thomson Brandt maintained a significant shareholding in this company (approximately 40%).
In 1982 both Thomson-Brandt and Thomson-CSF were nationalized by François Mitterrand. Thomson-Brandt was renamed Thomson SA (Société Anonyme) and merged with Thomson-CSF.
From 1983 to 1987 a major reorganisation of Thomson-CSF was undertaken, with divestitures to refocus the group on its core activities (electronics and defence). Thomson-CSF Téléphone and the medical division were sold to Alcatel and GE respectively. The semiconductor businesses of Thomson CSF was merged with Finmeccanica. Thomson acquired General Electric’s RCA and GE consumer electronics business in 1987.
In 1988 Thomson Consumer Electronics was formed, renamed Thomson Multimedia in 1995. The French government split the consumer electronics and defence businesses prior to privatisation in 1999, those companies being Thomson Multimedia (today Technicolor SA) and Thomson-CSF (today Thales Group).



Thomson-CSF was a major electronics and defence contractor. In December 2000 it was renamed Thales Group.



Thomson-CSF independence

Following the privatisation of the Thomson Group Thomson-CSF explored the possibility of merging with Marconi Electronic Systems, however British Aerospace was successful in that aim, forming BAE Systems.
In 2000 Thomson-CSF went through a series of transactions, including with Marconi plc. The major acquisition at this time was the £1.3 billion purchase of the British defence electronics firm, Racal. This made Thomson-CSF the second largest participant in the UK defence industry after BAE. Racal was renamed Thomson-CSF Racal plc.
On December 6, 2000 the group was renamed Thales.

------------------------------------------------------

NORDMENDE (THOMSON) CONTURA 63 (ICC9) CHASSIS ICC9 INTERNAL VIEW.





 
 
 

 







The NORDMENDE (THOMSON) CONTURA 63 (ICC9)  CHASSIS ICC9  is a sophisticated chassis which was a unusual engineering technology for a color tv. It doesn't reckon the more standardized schematic reference for a tv. The structure is more oriented in multifunction distributed chipset.






NORDMENDE (THOMSON) CONTURA 63 (ICC9)  CHASSIS ICC9 Switched mode power supply transformer


A switched mode power supply transformer, particularly for a television receiver, including a primary winding and a secondary winding with the primary winding and the secondary winding each being subdivided into a plurality of respective partial windings. The partial windings of the primary lie in a first group of chambers and the partial windings of the secondary lie in a second group of chambers of a chamber coil body, and the chambers of both groups are nested or interleaved with one another.







1. A switched mode power supply transformer, particularly for a television receiver, comprising in combination:
a primary winding and a secondary winding, with said primary winding being subdivided into three partial windings and said secondary winding being subdivided into two partial windings;
a chamber coil body having a plurality of chambers;
said partial windings of said primary winding being disposed only in a first group of said chambers, and said partial windings of said secondary winding being disposed only in a second group of said chambers, with each of said partial windings being disposed in a respective one of said chambers;
said chambers of said first group being interleaved with said chambers of said second group such that they alternate in sequence with said primary partial windings and said secondary partial windings being alternatingly disposed in five successive said chambers, so as to generate the major operating voltage at said secondary winding;
an additional secondary winding for generating a further operating voltage, said additional secondary winding likewise being subdivided into a plurality of partial windings; and,
said partial windings of said additional secondary winding are disposed only in respective said chambers of said second group below any of said partial windings of said secondary winding.


2. A transformer as defined in claim 1 wherein the total number of said chambers is six.

3. A transformer as defined in claim 1 wherein the width of the narrowest of said chambers is approximately 1 mm.

4. A transformer as defined in claim 1 or 2 wherein the widths of said chambers are different.

5. A transformer as defined in claim 1 or 2 wherein the total width of all of said chambers is only approximately 20 mm, whereby a flat and optimally coupled transformer is realized.

6. A transformer as defined in claim 1 wherein said additional secondary winding provides an operating voltage for a load which has a fluctuating current input.

7. A transformer as defined in claim 1 wherein said partial windings of said additional secondary winding are connected in parallel.

8. A transformer as defined in claim 1 wherein said partial windings of said primary winding are connected in series.

9. A transformer as defined in claim 1 or 8 wherein said partial windings of said secondary winding are connected in series.

10. A transformer as defined in claim 1 further comprising a plurality of auxiliary primary windings disposed in one chamber of said first group which is disposed in approximately the center of said first group and above the said partial winding of said primary winding disposed in said one chamber of said first group.

11. A transformer as defined in claim 1 wherein all of said partial winding disposed in said chambers of both said groups are wound with wire having the same diameter.

12. A switched mode power supply transformer as defined in claim 1 or 10 wherein: said coil body has six of of said chambers; said additional secondary winding is subdivided into three said partial windings; and two of said partial windings of said additional secondary winding are disposed below respective ones of said partial windings of said secondary winding and the third said partial winding of said additional secondary winding is disposed in the sixth said chamber.

13. A switched mode power supply transformer as defined in claim 10 further comprising at least one further secondary winding disposed in one of said chambers of said second group above any partial secondary winding present in said one of said chambers.

14. A switched mode power supply transformer, particularly for a television receiver, comprising in combination:
a primary winding and a secondary winding, with said primary winding and said secondary winding each being subdivided into a plurality of partial windings;
a chamber coil body having a plurality of chambers;
said partial windings of said primary winding being disposed only in a first group of said chambers, and said partial windings of said secondary winding being disposed only in a second group of said chambers with each of said partial windings being disposed in a respective one of said chambers;
said chambers of said first group being interleaved with said chambers of said second group such that said primary partial windings and said secondary partial windings are alternatingly disposed in successive said chambers, so as to generate the major operating voltage at said secondary winding;
an additional secondary winding for generating a further operating voltage, said additional secondary winding likewise being subdivided into a plurality of partial windings, and said partial windings of said additional secondary winding are disposed only in respective said chambers of said second group below any of said partial windings of said secondary winding.


15. A switched mode power supply transformer as defined in claim 1 or 14 wherein each of said partial windings of said primary winding contains the same number of turns and each of said partial windings of said secondary winding contains the same number of turns.

Description:
BACKGROUND OF THE INVENTION
The present invention relates to a switched mode power supply transformer, particularly for a television receiver.
In communications transmissions devices, particularly in television receivers, it is known to effect the desired dc decoupling from the mains by means of so-called switched mode power supply transformers. Such switched mode power supply transformers are substantially smaller and lighter in weight than a mains transformer for the same power operating at 50 Hz, because they operate at a significantly higher frequency of about 20-30 kHz. Such a switched mode power supply transformer (hereinafter called SMPS transformer) generally includes a primary side with a primary winding serving as the operating winding for the switch and further additional auxiliary windings, as well as a secondary side with a secondary winding for generating the essential operating voltage and possibly further additional windings for generating further operating voltages of different magnitude and polarity. The secondary and primary are insulated from one another as prescribed by VDE and have the necessary dielectric strength so that there is no danger of contact between voltage carrying parts on the secondary. A switched mode power supply (SMPS) circuit for a tv-receiver is described in U.S. Pat. No. 3,967,182, issued June 29, 1976.
A further requirement placed on such an SMPS transformer is that the stray inductance at least of the primary winding and of the secondary winding should be as small as possible. With too high a stray inductance, a transient behavior may develop during the switching operation which would not assure optimum switch operation of the switching transistor connected to the primary winding and would endanger this transistor by taking on too much power. Moreover, an increased stray inductance undesirably increases the internal resistance of the voltage sources for the individual operating voltages.
It is known to design the windings for such transformers as layered windings. Such layered windings, however, contain feathered intermediate foil layers and, after manufacture, generally require that the coil or the complete transformer be encased in order to insure VDE safety. Use as a chamber winding in television receivers presently does not take place because of the problems to be discussed below. A chamber winding would have the particular advantage that it could be wound more easily and economically by automatic machines. when using a chamber winding for a switched mode power supply, the detailed insulation between the primary and the secondary would be realized initially by two chambers with one of these chambers being filled only with the windings of the primary and the other of these chambers being filled only with the windings of the secondary. However, with such an arrangement there would exist only slight coupling between the primary and the secondary and thus an undesirably high stray inductance. If, on the other hand, the number of chambers were selected to be substantially larger, the transformer becomes more expensive and unnecessarily large. Moreover, a larger core would be required. Consequently, in the past, no television receiver has been introduced that included an SMPS transformer.
SUMMARY OF THE INVENTION
It is the object of the present invention to provide an SMPS transformer designed in the chamber wound technique which permits economical automatic winding, i.e. can be wound with but a single type of wire, has a structure which is spatially narrow and as flat as possible, provides the required insulation between the primary and secondary windings, and has a low stray inductance. The transformer should not be encased or saturated and nevertheless should produce no interfering noise during operation. The transformer should be able to be held in a circuit board without mechanical aids merely by its connecting terminals which are soldered to the circuit board.
The above object is basically achieved according to the present invention in that the transformer for a switched mode power supply, particularly for a television receiver, comprises: a primary winding and a secondary winding with the primary and secondary windings each being subdivided into a plurality of partial windings; and a chamber coil body with a plurality of chambers; and wherein the partial windings of the primary winding are disposed in a first group of chambers of the coil body, the partial windings of the secondary winding are disposed in a second group of chambers of the coil body, and the chambers of the first and second groups are interleaved.
Due to the fact that the individual windings or partial windings of the primary are disposed only in chambers of the first group and the windings or partial windings of the secondary are disposed only in chambers of the second group, i.e. primary and secondary are distributed to separate chambers, the necessary dielectric strength between primary and secondary is assured. By dividing each of the primary and secondary windings to a respective plurality or group of chambers and, due to the interleaved or nested arrangement of the chambers of the primary and the secondary, the desired fixed coupling between primary and secondary, and thus the desired low stray inductance at the primary and secondary, are realized. It has been found that a total number of chambers in the order of magnitude of six constitutes an economically favorable solution. With a smaller number of chambers, the coupling between primary and secondary is reduced. With a larger number of chambers, however, either the individual chambers become too small or the entire transformer, and particularly the core, become too large.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a circuit diagram for a preferred embodiment of a switched mode power supply transformer according to the invention.
FIG. 2 is a schematic partial sectional view showing the distribution of the individual windings of FIG. 1 to different chambers according to the invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
FIG. 1 shows a transformer intended for a switched mode power supply for a television receiver with a power output between 40 and 150 watts. The transformer includes a primary side P and a secondary side S which, while maintaining the required dielectric strength of, for example, 10,000 V, are galvanically decoupled or separated from one another. The primary side P includes a primary winding 1 which, as the operating winding, will lie in the collector circuit of a switching transistor switched at about 20-30 kHz. The primary winding 1 is divided into three partial windings 1a, 1b and 1c which are connected in series. When utilized in a television receiver, the beginning of partial winding 1a and the end of partial winding 1c are connected into the collector circuit of the switching transistor, while the taps between the partial windings 1a-1b and 1b-1c are not utilized, but rather form supporting points for the connection of the terminals of the partial windings. The primary side P also includes an additional winding 3 which feeds the feedback path with which the primary winding 1a-1c is designed as a self-resonant circuit. Moreover, the primary side P includes an additional winding 4 for regulating the moment of current flow in the switching transistor in the sense of stabilizing the amplitude of the output voltages on the secondary side S.
The secondary side S initially includes the secondary winding 2 from which is obtained, via a rectifier circuit (not shown), the main operating voltage U1. The secondary winding 2 is divided into two series connected partial windings 2a and 2b. Additionally, the secondary winding S includes a winding 5 for generating an operating voltage for the video amplifier and a further winding 6 for generating the operating voltage for the vertical deflection stage of a television receiver. Moreover, an additional secondary winding 7 is provided from which, after rectification, the operating voltage or the audio output stage of the receiver is obtained. Winding 7 comprises three partial windings 7a, 7b, 7c which are connected in parallel. The audio output stage of a television receiver has a greatly fluctuating current input between 50 mA and 1000 mA so that the load of the secondary side S varies considerably. This variation in load may effect an undesirable change in the operating voltage U1 which also influences the horizontal deflection amplitude. This undesirable dependency can be reduced in that the coupling between winding 7 and winding 4 is dimensioned greater, for regulating purposes, than the coupling between winding 2 and winding 4. This solution is described in greater detail in Federal Republic of Germany Offenlegungsschrift (laid open application) DE-OS No. 2,749,847 of May 10, 1979. This increased coupling between windings 7 and 4 is realized in the present case by the three parallel connected windings 7a, 7b, 7c. Finally, the secondary S includes a further winding 8 which serves to generate, after rectification, a negative operating voltage of -30 V.
FIG. 2 shows one half of the chamber coil body 9 for the individual windings of FIG. 1, with the body 9 including a total of six chambers 10. The size and particularly the widths of the individual chambers 10 can vary with respect to one another and the widths may all be different. Preferably, the width of the narrowest chamber 10 is about 1 mm and the total width of all six chambers is only approximately 20 mm so as to realize a flat and optimally coupled transformer.
As shown, one third of the primary winding 1, in the form of respective partial windings 1a, 1b and 1c, is distributed to each of the first, third and fifth chambers 10 of the coil body 9. The additional primary windings 3 and 4 are disposed in the third chamber 10 above the partial winding 1b. One half of the secondary winding 2, in the form of respective partial windings 2a, 2b, is distributed to each of the second and fourth chambers 10 of the coil body 9. The three partial windings 7a, 7b and 7c of the additional secondary winding 7 for the audio output stage are distributed to the second, fourth and sixth chambers 10, respectively, with the partial windings 7a-7c being disposed closest to the longitudinal axis of the coil body 9 and thus below any partial secondary winding 2a, 2b or other secondary winding which may be located in the same chamber. That is, the partial windings 7a and 7b are disposed below the partial windings 2a and 2b, respectively, in the respective second and fourth chambers 10, and below the additional secondary windings 5 and 8 in the sixth chamber 10. Further winding 6 is disposed above partial secondary winding 2b.
As can be seen in FIG. 2, the chambers 10 contain alternatingly only windings or partial windings of the primary side P or of the secondary side S. The illustrated nesting or interleaving of the windings, i.e. the alternating arrangement of windings of the primary side P and of the secondary side S in successive chambers 10, assures the desired close coupling between the primary side P and the secondary side S. The arrangement of the windings 3, 4 in approximately the center of the coil body 9 above partial winding 1b assures the desired close coupling between the windings 3, 4 with the other windings.
In an embodiment of the transformer shown in FIGS. 1 and 2 which was successfully tested in practice, the individual windings were all wound with the same diameter wire and contained the following numbers of turns:
______________________________________
Winding No. Number of Turns
______________________________________


1a 22

1b 22

1c 22

2a 30

2b 30

3 3

4 10

5 25

6 1

7a 11

7b 11

7c 11

8 16

______________________________________
The diameter of the wire of the windings 1-8 may be about 0.40 or 0.45 mm. Also, each winding may exist of two parallel shunted wires each of 0.3 mm diameter. The width of the six chambers 10--seen from the left to the right in FIG. 2--may be 0.95/1.95/1.75/1.95/0.95/2.75 mm and the thickness of the walls forming the chambers 0.65 mm.
It will be understood that the above description of the present invention is susceptible to various modifications, changes and adaptations, and the same are intended to be comprehended within the meaning and range of equivalents of the appended claims.


NORDMENDE (THOMSON) CONTURA 63 (ICC9)  CHASSIS ICC9  Raster distortion correction circuitry Unit for a video display apparatus that includes a square-planar picture tube

A video display apparatus incorporates a picture tube, such as a square-planar picture tube, that includes a phosphor screen located on a faceplate having an aspherically curved geometry that produces a relatively flat surface contour. Deflection circuitry generates line and field scanning currents in respective line and field deflection windings to enable the electron beams of the picture tube to scan a raster on the phosphor screen. The electron beams are subject to a scanning error that results in raster distortion, such as side pincushion distortion. A parabola generator produces a parabolic modulation of the scanning of the electron beams to generally correct the raster distortion while retaining a residual error due to the aspherically curved geometry of the faceplate. A nonlinear waveshaping circuit modifies the parabolic modulation of the scanning current in accordance with the aspherically curved geometry to provide additional modulation of the electron beam scanning for correcting the residual error. The nonlinear waveshaping circuit includes a current source and a switched current divider that is responsive to a threshold level of the parabolic voltage developed by the parabola generator.
This version is realized with the feature of an IC, principles of the means are almost the same.



1. Video display apparatus with correction of raster distortion, comprising:
a picture tube having a phosphor screen located on a faceplate having an aspherically curved surface contour, with at least one of a minor and major axis exhibiting a curvature that changes from an edge of the faceplate to the center;
deflection means including horizontal and vertical deflection windings for generating respectively therein horizontal and vertical deflection currents to enable an electron beam of said picture tube to scan a raster on said phosphor screen;
a parabola generator coupled to said deflection means for generating a generally parabolically shaped signal that produces a parabolic modulation of the scanning of said electron beam to generally provide correction of a first raster distortion while leaving uncorrected a residual raster distortion due to the change in curvature of said faceplate from said edge to the center; and
means for nonlinearly modifying said (parabolic modulation) parabolically shaped signal as a function of said change in curvature of said faceplate to provide additional modulation of the scanning of said electron beam for correcting said residual raster distortion.
2. Apparatus according to claim 1 wherein said first raster distortion results in side pincushion raster distortion and wherein said faceplate has an aspherically curved geometry that results in areas of said faceplate near the top and bottom being of increased curvature relative to the curvature in areas near the center. 3. Apparatus according to claim 2 wherein said residual raster distortion comprises a barrel-like distortion of a raster display of a vertical line pattern in said areas near the top and bottom of said faceplate. 4. Apparatus according to claim 2 wherein said nonlinearly modifying means comprises a current source coupled to said parabola generator and a switched current divider that switches from a first conductive state to a second conductive state during that portion of a vertical trace interval when a raster is being scanned on said faceplate in said areas of increased curvature. 5. Apparatus according to claim 4 wherein said parabolically shaped signal is a generally parabolically shaped input voltage repeating at a vertical deflection rate and having both a DC component voltage and an AC parabolic component voltage and wherein said switched current divider comprises an impedance receiving a constant current from said current source for establishing a bias voltage level and switching means responsive to said bias voltage level and direct current coupled to said parabola generator for shunting a portion of said constant current when said input voltage goes beyond a threshold voltage level established in accordance with said bias voltage level to modify the waveshape of said input voltage in a manner that corrects said residual raster distortion. 6. A circuit for correcting a given error of electron beam positioning in a video display apparatus, comprising:
a deflection winding;
an output stage for generating current in said deflection winding to control said electron beam positioning;
a parabolic voltage generator for developing a generally parabolically shaped input voltage repeating at a deflection rate and having both a DC component voltage and an AC parabolic component voltage;
an amplifier for driving said output stage in accordance with said parabolically shaped input voltage to generally correct said given error while retaining a residual error of said electron beam positioning;
a current source;
an impedance receiving current from said current source for establishing a bias voltage level; and
switching means responsive to said bias voltage level and direct current coupled to said parabolically shaped input voltage for shunting a portion of the current from said current source when said input voltage goes beyond a threshold voltage level established in accordance with said bias voltage level to modify the waveshape of said input voltage in a manner that corrects said residual error.
7. A circuit according to claim 6 wherein said switching means is direct current coupled to an output terminal of said parabolic voltage generator, at which terminal said input voltage is developed, and which is direct current coupled to a terminal of said impedance, at which impedance terminal said bias voltage level is developed. 8. A circuit according to claim 7 wherein said switching means comprises a diode in which there flows the shunted portion of current from said current source. 9. A circuit according to claim 7 wherein the shunted portion of current from said current source establishes a modified parabola voltage at an output terminal of said switching means that is direct current coupled to the output terminal of said parabolic voltage generator, said modified parabola voltage having a waveform that generally follows the waveform of said input voltage when said switching means is in one conductive state and having a waveform of waveshape that is substantially different than that of said input voltage when said switching means is in another conductive state. 10. A circuit according to claim 9 including means for AC coupling said modified parabola voltage to said amplifier. 11. A circuit according to claim 10 including means for adjusting said bias voltage level to adjust the switching instants of said switching means and wherein a peak amplitude of said modified parabola voltage remains substantially unchanged for different adjustments of said bias voltage level. 12. A circuit according to claim 11 including an adjustable voltage divider coupled to said AC coupling means and direct current coupled to said amplifier for adjusting the amplitude of the AC coupled modified parabola voltage, and wherein the peak amplitude of the modified parabola voltage that is established at the output terminal of said switching means remains substantially unchanged for different adjustments of said voltage divider. 13. A circuit according to claim 9 wherein said input voltage includes an AC sawtooth component voltage repeating at said deflection rate and further including means for applying said sawtooth component voltage to said impedance to provide common-mode rejection of said sawtooth component voltage with respect to the shunted portion of current from said current source. 14. A circuit according to claim 9 wherein said switching means applies an additional voltage to said terminal of said impedance that is representative of the difference between said bias voltage level and said input voltage to control the amplitude of the shunted portion of current in accordance with said additional voltage. 15. A circuit according to claim 9 wherein said current source comprises a source of DC voltage coupled to a second impedance, and wherein said switching means applies an additional voltage to said terminal of the first mentioned impedance that is representative of a difference between said bias voltage level and said input voltage. 16. A circuit according to claim 15 wherein said DC voltage of said DC voltage source is substantially greater in magnitude than that of said additional voltage to prevent significant changes from occurring in the magnitude of said current source when said switching means changes conductive states. 17. A circuit according to claim 6 wherein said video display apparatus includes a picture tube having a phosphor screen located on a faceplate having an aspherically curved geometry that produces a relatively flat surface contour and wherein said residual error of electron beam positioning is produced in accordance with said aspherically curved geometry. 18. A circuit according to claim 17 wherein said given error is produced by side pincushion distortion and wherein said residual error produces a barrel-like distortion of a raster display of a vertical line pattern in regions near the top and bottom of said faceplate where the curvature of said faceplate is increased relative to the curvature in the center region. 19. Apparatus according to claim 1 wherein said nonlinearly modifying means produces a parabolic modulation when scanning raster lines near top and bottom of said raster that is different than when scanning raster lines near raster center for correcting said residual raster distortion due to the aspherically curved geometry of said faceplate.
Description:
This invention relates to electron beam positioning error correction circuitry in a video display apparatus.
New, flatter faceplate picture tubes, such as the RCA Corporation square-planar picture tubes, have aspherically curved faceplate contours. A tube of this type, having a complex curvature faceplate, is described in the following U.S. patent applications, herein incorporated by reference.
1. U.S. patent application Ser. No. 469,772, filed Feb. 25, 1983, by F. R. Ragland, Jr. entitled CATHODE-RAY TUBE HAVING AN IMPROVED SHADOW MASK CONTOUR.
2. U.S. patent application Ser. No. 469,774, filed Feb. 25, 1983 by F. R. Ragland, Jr. entitled CATHODE-RAY TUBE HAVING A FACEPLATE PANEL WITH A SUBSTANTIALLY PLANAR PERIPHARY.
3. U.S. patent application Ser. No. 469,775, filed Feb. 25, 1983, by R. J. D'Amato et al., entitled CATHODE-RAY TUBE HAVING DIFFERENT CURVATURE ALONG MAJOR AND MINOR AXES.
4. U.S. patent application Ser. No. 529,644, filed Sept. 6, 1983, by R. J. D'Amato et al., entitled CATHODE-RAY TUBE HAVING A FACEPLATE PANEL WITH AN ESSENTIALLY PLANAR SCREEN PERIPHERY.
In one form of flatter faceplate picture tube, as typified by the RCA 110° COTY-SP, square-planar, 27 V, color television picture tube, A68ACC10X, the formula for the tube faceplate sagittal height, z, in millimeters, with reference to the center of the faceplate is given by: Z=A 1 X 2 +A 2 X 4 +A 3 Y 2 +A 4 X 2 Y 2 +A 5 X 4 Y 2 +A 6 Y 4 +A 7 X 2 Y 4 +A 8 X 4 Y 4 ,
where X and Y are the distance coordinates, in millimeters, from the faceplate center along the major and minor axes, respectively, and where: A 1 =-0.236424229×10 -4 A 2 =-0.363538575×10 -8 A 3 =-0.422441063×10 -3 A 4 =-0.213537355×10 -8 A 5 =+0.883912220×10 -13 A 6 =-0.100020398×10 -9 A 7 =+0.117915353×10 -14 A 8 =+0.527722295×10 -21
The picture tube faceplate defined by this formula has a relatively shallow curvature near the center of the faceplate, which increases near the edges along paths parallel to both the major and minor axes of the tube. The overall result is a faceplate of relatively flat appearance and with planar edges, namely, with points along the top, bottom, right and left edges located substantially in a common plane.
In general, the raster scanned on the phosphor screen of a picture tube may exhibit an east-west or side pincushion raster distortion. This distortion may be substantially corrected by a side pincushion correction circuit that produces a parabolic amplitude modulation of the horizontal deflection current. The required vertical rate parabola voltage may be obtained from the vertical deflection circuit by integration of the vertical sawtooth current.
When using such a conventional circuit for raster correction in a picture tube, such as a square-planar picture tube that has an aspherically curved faceplate, a small but often objectionable raster distortion may remain at the top and bottom of the display, as illustrated by the solid vertical lines of a raster R display R in FIG. 1. Due to the aspherically curved geometry of the faceplate, the vertical line bend away from the vertical direction near the top and bottom of the raster display, where the curvature of the faceplate increases. The overall appearance of the vertical lines in FIG. 1 is slightly barrel-shaped.
In accordance with an aspect of the invention, a video display apparatus includes a picture tube having a phosphor screen located on a faceplate having an aspherically curved geometry. Correction circuitry is provided that compensates for raster distortion or electron beam positioning errors, such as pincushion or convergence distortions. The correction circuitry includes a parabola generator coupled to deflection circuitry for producing a parabolic modulation of the scanning of the electron beam. The modulation generally provides correction of the raster distortion or electron beam positioning error, while retaining a residual error due to the aspherically curved geometry of the faceplate. The modulation is nonlinearly modified in accordance with the aspherically curved geometry to provide additional modulation of the scanning of the electron beam for correcting the residual error.
In accordance with another aspect of the invention, a particularly advantageous arrangement of correction circuitry smoothly straightens the bent vertical lines near the top and bottom of the raster display illustrated in FIG. 1.
The parabola generator of the correction circuit arrangement produces a generally parabolically shaped input voltage repeating at a deflection rate. An amplifier drives an output stage that generates current in a deflection winding to control the electron beam landing position. The amplifier is responsive to the output of the parabola generator for driving the output stage in accordance with the parabolically shaped input voltage to generally correct electron beam landing error, while retaining a residual error. A current source with a switched current dividing network is responsive to a threshold level of the parabolically shaped input voltage for supplying to the amplifier a portion of the current from the current source when the input voltage exceeds the threshold level to correct the residual error.
FIG. 1 illustrates vertical lines of a raster display scanned on the screen of a square-planar picture tube;
FIG. 2 illustrates in block diagram form raster scanning circuitry used in conjunction with raster scanning on the aspherically curved faceplate of a square-planar picture tube, shown schematically in FIG. 2 from a side elevation view, in partial breakaway;
FIG. 3 illustrates detailed embodiments of the horizontal and vertical deflection circuits of FIG. 2, including circuitry embodying the invention that corrects for the residual side pincushion distortion illustrated in FIG. 1;
FIG. 4 illustrates a detailed embodiment of a portion of the circuit of FIG. 3; and
FIG. 5 illustrates waveforms useful in explaining operation of the circuits of FIGS. 3 and 4.
In FIG. 1, there is illustrated vertical lines of a raster display R that is generated on the phosphor screen of a faceplate 30 of a square-planar picture tube SP of FIG. 2. Horizontal and vertical deflection circuits 20 and 40 of FIG. 2 generate horizontal and vertical deflection currents in horizontal and vertical deflection windings L H and L V , respectively. The horizontal and vertical deflection currents deflect electron beams 18 in square-planar picture tube SP to produce raster display R on faceplate 30.
Square-planar picture tube SP incorporates a glass envelope 11 comprising a generally rectangular faceplate panel 19 and a tubular neck 14 connected by a funnel 16. Panel 19 comprises viewing faceplate 30 and a peripheral flange or side wall 12, which is sealed to funnel 16 by a glass frit 17. A generally rectangular 3-color cathodoluminescent phosphor screen 15 is carried by the inner surface of faceplate 30. The screen may be a line screen, with the phosphor lines extending substantially parallel to the minor or vertical axis Y--Y of the tube. Alternatively, the screen may be a dot screen. A multi-aperture color selection electrode or shadow mask 13 is removably mounted within panel 19 in predetermined spaced relation to screen 15. An electron gun 10, shown schematically by dashed lines in FIG. 2, is centrally mounted within neck 14 to generate and accelerate the three electron beams 18 along convergent paths through mask 13 to screen 15.
The line and field deflection currents in line and field deflection windings L H and L V , respectively, subject the three electron beams 18 to vertical and horizontal magnetic flux that scans the beams horizontally in the direction of the major or horizontal axis X--X and vertically in the direction of the minor axis Y--Y, in a rectangular raster pattern over screen 15. The longitudinal axis of picture tube SP is labeled Z--Z in FIG. 2.
Faceplate 30 of square-planar color picture tube SP is relatively flat. The curvature of the faceplate is complex and may be approximated in accordance with the polynominal expression given above. In the field scanning direction, or as the electron beams are scanned from top edge to bottom edge, vertically, the curvature of the faceplate decreases from top edge to center and then increases again to the bottom edge. A similar situation holds in the line scanning direction.
Assume that horizontal and vertical deflection circuits 20 and 40 of FIG. 2 correct for distortions such as S-distortion, north-south and gullwing distortion. Furthermore, assume that horizontal deflection circuit 20 generally corrects for side pincushion distortion by parabolically modulating the amplitude of the horizontal deflection current. The display of a vertical line pattern on faceplate 30 by means of raster line scanning produces a raster display such as illustrated in solid-line in FIG. 1. The generally vertical lines exhibit a residual distortion at the top and bottom of the raster which, due to the aspherically curved geometry of faceplate 30, causes the vertical raster lines to be bent inward toward the center in a barrel-like manner.
In accordance with a feature of the invention, horizontal deflection circuit 20 nonlinearly modifies the parabolic modulation of the horizontal deflection current to straighten the vertical lines at the top and bottom of the raster, as illustrated by the dashed-line, straight line segments of FIG. 1.
FIG. 3 illustrates detailed embodiments of vertical deflection circuit 40 and horizontal deflection circuit 20 of FIG. 2 that include nonlinear waveshaping circuitry in accordance with an aspect of the invention. In horizontal deflection circuit 20 of FIG. 3, a B+ voltage is applied to the primary winding W p of a flyback transformer T1 via a small valued resistor 21. A capacitor 22 provides filtering. Primary winding W p is coupled to a horizontal output stage 70 of horizontal deflection circuit 20. Horizontal output stage 70 includes a horizontal oscillator and driver 25, a horizontal output transistor Q1, a first retrace capacitor C R1 coupled across transistor Q1, a damper diode D2, a second retrace capacitor C R2 coupled across diode D2 and the series arrangement of a linearity inductor 26, and S-shaping capacitor C s , deflection winding L H of FIG. 2, and a resonant circuit 27, comprising a capacitor C1 in parallel with the inductance of a tapped winding W1 of a transformer T2. Deflection winding L H is coupled to the tap terminal of winding W1.
Resonant circuit 27 is tuned to produce approximately two cycles of oscillation during the horizontal trace interval for introducing an oscillatory current component into horizontal deflection current i H that provides dynamic S-correction of the raster scanned on faceplate 30 of square-planar picture tube SP of FIG. 1. The function of dynamic S-correction, resonant circuit 27 is more fully described in U.S. Pat. No. 4,563,618, by P. E. Haferl, issued Jan. 7, 1986, entitled S-CORRECTED DEFLECTION CIRCUIT.
To provide side pincushion correction, the amplitude of horizontal deflection current i H is modulated at a vertical rate by a side pincushion correction modulator circuit 30 that drives horizontal output circuit 70. Side pincushion correction circuit 30 includes a damper diode D3 with a grounded anode and a cathode coupled to the anode of damper diode D2, a retrace capacitor C R3 coupled across diode D3, a modulator choke inductor L m coupled to the junction of retrace capacitors C R2 and C R3 , and a modulator control circuit 60 coupled to choke L m .
Modulator control circuit 60 modulates at a vertical rate the modulator current i m in choke L m , to concurrently modulate the retrace pulse voltage V Rm developed across modulator retrace capacitor C R3 . The modulation of retrace pulse voltage V Rm produces a concurrent but opposite sense modulation of deflection retrace pulse voltage V Rd across deflection retrace capacitor C R2 . The vertical rate modulation of deflection retrace pulse voltage V Rd produces the required vertical rate modulation of horizontal deflection current i H that provides side pincushion correction.
The opposing sense modulation of retrace pulse voltages V Rd and V Rm produces an unmodulated retrace pulse voltage V R at the collector of horizontal output transistor Q1. Retrace pulse voltage V R is applied to the primary winding W p of flyback transformer T1 for generating an unmodulated retrace pulse voltage V RH at a terminal A of a secondary winding W s . An unmodulated retrace pulse voltage is also generated in a high voltage winding W HV for developing an ultor accelerating potential at a terminal U of a high voltage generating circuit 23.
FIG. 3 also illustrates a detailed embodiment of vertical deflection circuit 40 of FIG. 2. Vertical deflection circuit 40 includes a vertical deflection amplifier U1 coupled to vertical deflection winding L V of FIG. 2 for generating a vertical deflection current i V that deflects the electron beams in picture tube SP of FIG. 2 in the vertical direction. Vertical deflection winding L V is coupled to a north-south and gullwing distortion correction circuit 34 for modulating vertical deflection current i V in a manner that corrects both north-south pincushion distortion and gullwing distortion of the raster when scanning raster lines on square-planar picture tube SP of FIG. 2. A description of the operation of north-south pincushion and gullwing correction circuit 34 may be found in U.S. patent application Ser. No. 719,227, filed Apr. 2, 1985, by P. E. Haferl entitled NORTH-SOUTH PINCUSHION CORRECTED DEFLECTION CIRCUIT, now U.S. Pat. No. 4,668,897 and in U.S. patent application Ser. No. 733,661, filed May 10, 1985, by P. E. Haferl et al, entitled GULLWING DISTORTION CORRECTED DEFLECTION CIRCUITRY FOR A SQUARE-PLANAR PICTURE TUBE, both herein incorporated by reference.
Vertical deflection current i V , after passing through correction circuit 34, flows through a coupling or vertical S-shaping capacitor C V and a current sampling resistor R s . Coupling capacitor C V intergrates vertical deflection current i V to develop across the capacitor between terminals 35 and 36 an AC parabola voltage, of almost ideal waveshape, that repeats at the vertical deflection rate. The voltage across capacitor C V comprises the AC parabola voltage superimposed upon a DC level established by vertical deflection amplifier U1. The voltage V sV developed across current sampling resistor R s is illustrated in FIG. 5a and comprises an AC, S-shaped, sawtooth voltage repeating at the vertical deflection rate. The voltage V1 developed at terminal 35 is illustrated in FIG. 5b and equals the sum of the voltages developed across capacitor C V and resistor R s . Thus, the AC component of voltage V1 during the vertical trace interval t 3 -t 7 of FIG. 5b is a generally parabolically shaped voltage that is skewed downward by the relatively small sawtooth voltage component derived from sampling resistor R s .
The voltages at terminals 35 and 36 are applied to vertical deflection amplifier U1 to provide DC and AC feeback, respectively, to the amplifier. The vertical rate voltages V1 and V sV , the horizontal rate retrace pulse voltage V RH and a voltage V ds developed across winding W2 of transformer T2 are coupled to correction circuit 34 to provide deflection synchronization information and to provide waveform information that produces the required waveshaping and modulation of vertical deflection current i v , as described in the aforementioned U.S. patent applications.
Vertical deflection circuit 40 may be considered as a low impedance voltage source 48 that generates parabola voltage V1 at output terminal 35 of the source.
Vertical parabola voltage V1 is nonlinearly waveshaped by a nonlinear network 50 and is then applied via a DC blocking capacitor C2 and a parabola amplitude adjusting potentiometer R a to the noninverting input terminal of an amplifier U2 of side pincushion control circuit 60. Vertical sawtooth voltage V sV is applied to the inverting input terminal of amplifier U2 via the wiper arm of a trapeze adjusting potentiometer R t and a resistor 31. The DC level at the inverting input terminal is controlled by a width adjusting potentiometer 33 that couples a +25 V source to the inverting input terminal via a resistor 32 and the wiper arm of potentiometer 33. The output of amplifier U2 is coupled to an inverting driver stage U3 that applies a modulation voltage V m to modulator choke inductor L m .
Side pincushion control circuit 60 is operated in the switched mode at the horizontal rate. A horizontal sawtooth voltage generator 29, synchronized by horizontal retrace pulse voltage V RH , applies a horizontal rate sawtooth voltage V sH to the noninverting input terminal of amplifier U2 via a resistor R g . Resistor R g represents the effective source impedance of sawtooth voltage generator 29. The output of amplifier U2 is a pulse width modulated, horizontal rate voltage having a duty cycle that varies at a vertical rate. Modulation voltage V m therefore is also a pulse width modulated, horizontal rate voltage having a duty cycle that varies at a vertical rate. This enables the drive provided by side pincushion modulator circuit 30 to be varied in a manner that corrects side pincushion distortion.
Side pincushion modulator circuit 30 operates in a manner similar to that described in U.S. patent application Ser. No. 651,301, filed Sept. 17, 1984, now U.S. Pat. No. 4,634,937 by P. E. Haferl, entitled EAST-WEST CORRECTION CIRCUIT. Other side pincushion correction circuits, such as switched mode diodc modulator circuits, may be used to drive horizontal output stage 70.
In accordance with an aspect of the invention, nonlinear waveshaping network 50 is interposed between terminal 35 and the noninverting input terminal of amplifier U2 of side pincushion control circuit 60. Nonlinear network 50 modifies the waveshape of parabola voltage V1 at terminal 35 to generate a modified parabola voltage V2 at a terminal 37, as illustrated by the solid-line waveform of voltage V2 in FIG. 5f. Shaped parabola voltage V2 is then applied to the noninverting input terminal of amplifier U2 via AC coupling capacitor C2 and potentiometer R a . The additional waveshaping provided by nonlinear network 50 corrects the residual side pincushion error that would otherwise exist when scanning a raster on the phosphor screen of a square-planar picture tube.
Nonlinear waveshaping network 50 comprises a constant current source CS in series with a potentiometer R2 that is coupled to the wiper arm of trapeze adjusting potentiometer R t . A diode D1, functioning as a unidirectional switch, is coupled between the wiper arm of potentiometer R2 and terminal 37, with the cathode of diode D1 being coupled to terminal 37.
In operation, constant current source CS generates an almost ideal constant current i 0 , illustrated in FIG. 5d, that does not significantly change in value throughout the entire vertical deflection interval t 3 -t 8 . Diode D1 is reverse biased by voltage V2 during the interval t b of FIG. 5. During this interval, all of current i 0 that flows into end terminal 38 of potentiometer R2 flows out of the other end terminal 39, as illustrated in FIG. 5c by the current i 3 during the interval t b . Current i 3 flows in that portion R2b of potentiometer R2 between intermediate wiper arm terminal 41 and end terminal 39 coupled to the wiper arm of potentiometer R t . The solid-line waveform of FIG. 5c also illustrates the voltage V3 developed by current i 3 in resistance R2b.
When diode D1 is nonconductive, during the interval t b of FIG. 5, constant current source CS advantageously establishes an adjustable DC bias voltage level V b at intermediate wiper arm terminal 41, as illustrated by the dotted-line waveform of FIG. 5c. Voltage V b equals the constant voltage level V 0 that is established for voltage V3 by constant current source CS, summed with the vertical sawtooth voltage developed at the wiper arm of trapeze adjusting potentiometer R t . Illustratively, voltage level V 0 is shown in FIG. 5c at a level established by the wiper arm of potentiometer R2 when the wiper arm is in a centered position.
When diode D1 is nonconductive, voltage V1 is divided by a voltage divider (R1, R a , R b , R g ) coupled between terminal 35 and the noninverting input terminal of amplifier U2 for developing voltage V2 at terminal 37, which terminal is an intermediate point of the voltage divider. As illustrated in FIGS. 5b and 5f, voltages V1 and V2 exhibit substantially the same waveshape during the interval t b .
During the second half of vertical trace, after the center of trace instant t 5 , voltages V1 and V2 decrease in amplitude. Near time t 6 , voltage V1 has decreased to a threshold voltage level V' al and voltage V2 has decreased to a threshold voltage level V a1 . The decreased voltage V2 at terminal 37 near time t 6 or time t 1 , enables diode D1 of nonlinear waveshaping circuit 50 to begin conducting, thereby coupling together terminals 37 and 41.
Diode D1 continues to conduct throughout the interval t a of FIG. 5. During this interval, voltages V1 and V2 are below the threshold levels V' a2 and V a2 , respectively. At the end of the interval t a , near time t 9 or time t 4 , voltages V1 and V2 have increased sufficiently to reestablish at terminal 41 the bias voltage level V b of FIG. 5c that forces diode D1 to become nonconductive.
During the interval t a , when diode D1 is conductive, a portion of constant current i O of FIG. 5d, that flows in the upper resistance portion R2a of potentiometer R2, is shunted away from resistance R2b via the wiper arm of potentiometer R2 and diode D1. The shunt current i c in diode D1 is illustrated in FIG. 5e during the interval t 1 -t 4 or t 6 -t 9 . Current i c substracts from constant current i 0 when diode D1 is conductive to reduce the amplitude of current i 3 in resistance R2b by the amount of current shunted. As illustrated in FIG. 5c, current i 3 , during the interval t a , has the same waveshape as current i c of FIG. 5e, but inverted in phase.
The waveshape of current i c is determined in accordance with the waveshape of the parabolic component of voltage V1 that is applied to the voltage divider (R1, R2b) formed whendiode D1 is conductive. The amplitude of current i c is related to the difference in value between the bias voltage level V b and the parabolic voltage V1.
Correction current i c , flowing into terminal 37, modifies the waveshape of voltage V2 during the interval t a to correct the residual side pincushion error that would otherwise exist in raster display R of FIG. 1. Correction current i c flows mainly in resistor R1 to provide an additional voltage drop between terminals 37 and 35 that produces a flatter slope to the sides of parabola voltage V2 during the conduction interval t a of diode D1.
The solid-line waveform of FIG. 5f during the interval t a illustrates voltage V2 with diode D1 conducting. The dashed-line waveform illustrates the waveshape that voltage V2 would have assumed had diode D1 remained nonconductive during the interval t a . Comparing the solid-line waveform with the dashed-line waveform in FIG. 5f, one notes that the presence of nonlinear network 50 waveshapes parabola voltage V2 during the intervals t 3 -t 4 and t 6 -t 7 , when the top and bottom of the raster are being scanned.
The flattening of parabola voltage V2 occurs when the raster lines between lines L3 and L4 and between lines L6 and L7 of FIG. 1 are being scanned. This flattening produces less modulation of the amplitude of deflection current i H when scanning the top and bottom raster lines. The result of the nonlinear waveshaping is the straightening of the bent vertical line segments of raster display R of FIG. 1 to correct the residual side pincushion error caused by the increased curvature of the faceplate of a square-planar picture tube in the top and bottom regions of the faceplate.
FIG. 4 illustrates a more detailed embodiment of a portion of the circuitry of FIG. 3 that includes nonlinear network 50. Items in FIGS. 3 and 4 similarly identified perform similar functions or represent similar quantities.
Inverting driver stage U3 comprises a switching transistor Q2 driven at its base by amplifier U2 and having its collector coupled to choke inductor L m and its emitter coupled to ground. During those intervals within each horizontal deflection cycle that transistor Q2 is cutoff, modulator current i m flows to the B+ supply via a flywheel diode D4. DC biasing for transistor Q2 is established by voltage dividing resistors 42 and 43. To provide stabilized operation of driver transistor Q2, negative feedback from the collector of the transistor to the noninverting input terminal of amplifier U2 is provided via a resistor 44.
Horizontal sawtooth generator 29 comprises an RC network including a resistor 45 coupled to flyback transformer terminal A and a capacitor 46 coupled to the noninverting input terminal of amplifier U2. DC biasing of the noninverting input terminal is provided by resistor R b . Horizontal retrace pulse voltage V RH is integrated by the RC network to develop the horizontal sawtooth voltage V sH that produces the horizontal rate switching of transistor Q2. The duty cycle of the horizontal rate switching is modulated by means of the vertical rate modulation of the AC-zero level of voltage V sH .
In FIG. 4, constant current source CS comprises a DC voltage source of relatively large magnitude, such as the 140 volt, B+ voltage source, coupled to a resistor R3 of relatively large value, such as 180 kilohm. The amplitude of constant current i 0 is mainly determined by the value of the B+ voltage divided by the sum of the values of resistors R3 and R2. Current i 0 establishes an adjustable bias voltage level V b at the wiper arm of potentiometer R2 that maintains diode D1 nonconductive during the interval t b of FIG. 5 when parabola voltage V1 is sufficiently large in amplitude to keep the diode reverse biased. During the remaining interval t a , parabola voltage V1 is sufficiently small in amplitude to enable diode D1 to shunt some of current i 0 away from resistance portion R2b of potentiometer R2 to provide the correction current i c that waveshapes parabola voltage V2.
Advantageously, diode D1 is DC coupled to parabola voltage source 48 via resistor R1, with the cathode of diode D1 being coupled on the DC side (with respect to parabola voltage V1) of coupling capacitor C2. By means of the DC connection of diode D1 to parabola voltage source 48, the diode switching instants t 4 and T 6 may be adjusted by potentiometer R2 independently of the adjustment of east-west parabola amplitude potentiometer R a . When potentiometer R a is adjusted for the desired parabola amplitude, the AC-zero level of the parabola voltage applied to the noninverting input terminal of amplifier U2 also varies. This variation of the AC-zero level has little or no effect on the switching of diode D1.
For example, assume amplitude potentiometer R a is adjusted to provide the proper amount of side pincushion correction when the central raster lines are being scanned between raster line L4 and raster line L6 of FIG. 1 during the interval t 4 -t 6 of FIG. 5. Potentiometer R2 may then be adjusted to establish a bias voltage level V b that enables diode D1 to switch conductive states near times t 4 and t 6 . The switching of diode D1 near times t 4 and t 6 provides the required additional waveshaping of voltage V2 that corrects the residual side pincushion error in the top and bottom regions of raster display R.
The instants when diode D1 switches conductive states are controlled by the DC bias level V b established by potentiometer R2 rather than by amplitude potentiometer R a . The adjustment of potentiometer R2 has no significant effect on the previous amplitude adjustment provided by potentiometer R a .
When the wiper arm of potentiometer R2 is moved towards end terminal 39, the conduction interval t a of diode D1 decreases and the cutof interval t b increases. The location of raster lines L4 and L6 where nonlinear waveshaping begins moves away from center raster line L5 towards top and bottom raster lines L3 and L7, respectively. The peak downward excursion of voltage V2, that occurs near the beginning of retrace near times t 2 and t 7 , also moves downward toward the dashed-line level that represents the peak downward excursion when diode D1 is cutoff for the entire vertical deflection interval t 2 -t 7 .
The amount of nonlinear waveshaping of parabola voltage V2 may be defined as the voltage difference between the dashed and solid-line waveforms V2 of FIG. 5f at times t 3 and t 7 , the start and end of vertical trace, respectively. This voltage difference relative to the dashed-line waveform V2 represents the amount of correction resulting on the raster display of FIG. 1.
When the wiper arm of potentiometer R2 is moved toward end terminal 38, the amount of nonlinear waveshaping increases until the conduction interval t a of diode D1 equals the nonconduction interval t b . As the wiper arm of potentiometer R2 is moved further toward end terminal 38, the amount of waveshaping begins to decrease and reaches zero when bias voltage level V b is set at a sufficiently high level to enable diode D1 to conduct for the entire vertical deflection interval.
The amplitude of current i c changes when potentiometer R2 is adjusted. However, the peak amplitude that parabola voltage V2 attains remains substantially the same at all levels of adjustment because correction current i c of FIGS. 3 and 4 flows mainly in resistor R1 and does not add any significant charge to AC coupling capacitor C2. No significant portion of current i c flows in capacitor C2 because of the long time constant associated with capacitor C2 and resistors R a , R b , R g . Current i c causes a very small increase of the average DC voltage at terminal 37, not illustrated in waveform FIG. 5. This increase amounts to approximately 75 millivolt, which is 1/4 the voltage difference between the dashed and solid-line waveforms of voltage V2 in FIG. 5f, at times t 3 and t 7 .
When diode D1 becomes conductive, parabola voltage source 48 becomes DC coupled to current source CS. At the same time an additional load impedance becomes coupled to current source CS derived from the voltage divider (R1, R a , R b , R g ). Because of the DC connection provided by diode D1 between the wiper arm of potentiometer R2 and parabola voltage source 48, the additional voltage that is coupled in-circuit with current source CS during the interval t a of FIG. 5 is relatively small. The additional voltage, ΔV3, equals the voltage difference between the constant voltage level V 0 of FIG. 5c and the voltage V3 developed across resistance R2b.
The peak-to-peak amplitude of voltage ΔV3 is relatively small, approximately one volt peak-to-peak for the values given in FIG. 4. Because the peak-to-peak amplitude ΔV3 is much smaller than the B+ voltage of constant current source CS, the shunt current i c that source CS supplies when diode D1 is conductive is also small relative to current i 0 and has substantially no effect on the amplitude of the constant current. For the values given in FIG. 4, the amplitude of current i 0 changes less than one percent during the conduction interval t a shown in FIG. 5.
End terminal 39 of potentiometer R2 may be advantageously coupled to trapeze adjusting potentiometer R t rather than to ground. This connection enables nonlinear network 50 to provide a common-mode rejection of the sawtooth voltage component of parabola voltage V1. Thus, diode D1 nonlinearly waveshapes only the parabolic component of voltage V1 and not the sawtooth component. The common-mode rejection of the waveshaping of the sawtooth component of voltage V1 may be noted from the waveforms of FIGS. 5c and 5e, which are symmetrical about the center of trace instant t 5 .
Nonlinear waveshaping network 50 advantageously produces gradual changes in the slope of parabola voltage V2 at the switching instants of diode D1 that smoothly straightens the bent segments of the vertical lines of raster display R in FIG. 1, without introducing wiggly line excursions of the vertical line pattern near raster lines L4 and L6.
Nonlinear network 50 operates as a current divider to divide current i 0 into current i 3 and current i c during conduction of diode D1. Thus, the change in the forward voltage drop of diode D1, produced by variations in ambient temperature, has little influence on the waveshaping of the parabola voltage.



SGS-THOMSON MICROELECTONICS STV2161 VIDEO SCANNING PSI PROCESSOR:


The STV2161 isan I2ic bus controlled Video Scanning and PSI (Picture Signal Improvement) processor for 1H deflection (15.625kHz) television sets.
The IC incorporates also an secondary SMPS
(Switch Mode Power Supply) controller with soft-start and protection facilities.


VIDEO PART

I RGB INPUT WITH FAST BLANKING SWITCH
FOR SCART CONNECTOR

I RGB INPUT WITH FAST BLANKING FOR ON
SCREEN DISPLAY / TELETEXT

I USER CONTROL FUNCTIONS : CONTRAST,
BRIGHTNESS, SATURATION

I AVERAGE BEAM LIMITER FUNCTION WITH
CONTRAST AND BRIGHTNESS REDUC-
TION

I PAUSECAM OR NTSC MATRIX SWITCH-
ABLE BY BUS

I AUTOMATIC CUTOFF LOOP WITH INTE-
GRATED DIGITAL MEMORY AND LEAKAGE
CURRENT COMPENSATION

I CUTOFF REGISTER STATUS READABLE
BY BUS

I FULL WHITE (DRIVE) ADJUSTMENT VIA
BUS (NO CONTROL LOOP)

I BUS REPLY : FAST BLANKING SIGNAL
RECOGNITION FOR POSITIVE AND NEGA-
TIVE LEVEL/ TUBE COLD INFORMATION

I OVERSIZE BLANKING FOR VERTICAL AND
HORIZONTAL OVERSIZE

I APX (AUTO—P|X) : ADAPTIVE CONTRAST
CONTROL

I COLOR MATCHING (IN CONNECTION WITH
WHTTE STRETCH ; SEE PSI PART)

I APL (AUTO-PEDESTAL) ADAPTIVE
BRIGHTNESS (IN CONNECTION WITH
BLACK STRETCH; SEE PSI PART)

SCANNING PART

I SYNC SEPARATOR WITH BLACK LEVEL
CLAMP BASED ON CURRENT RATIO 1 :8

I PHI1 LOOP WITH COMPLETELY INTE-
GRATED CALIBRATED RC OSCILLATOR
(NO EXTERNAL CERAMIC) AND EXTERNAL
LOOP FILTER

I PHI1 PHASE POSITION SELECTABLE BY
BUS FOR COMB FILTER APPLICATION

I VCR AND TV MODE (SELECTION OF CURRENT) FOR PHI1 LOOP FILTER

April 1995

ADVANCE DATA

- AUTOMATIC SYNC PRESENCE DETECTION
AND AUTOMATIC PHI1 MUTE (FREE RUN
ING) MODE (IMPROVED COMPARED TO
STV2160)

- SYNCHRONISATION OUTPUT FOR VIDEO
TEXT AND OSD DISPLAY

- PH|2 LOOP WITH INTERNAL LOOP FILTER

- PULSE OUTPUT FOR DRIVING HORIZONTAL OUTPUT STAGE

- HORIZONTAL DRIVE STANDBY VOLTAGE
>VCc POSS|BLE(12V MAX)

- VERTICAL COUNTDOWN CIRCUIT FOR
CLEAN VERTICAL DEFLECTION

- INTEGRATED VERTICAL SAWTOOTH GENERATOR WITH AMPLITUDE CONTROL
LOOP

- INTERLACE MODE CONTROLLED BY BUS

- VERTICAL SIZE CORRECTION (BREATHING) TO ADAPT DEFLECTION SENSITIVITY
TO THE CURRENT BEAM (HIGH VOLTAGE VARIATION)

- BUS ADJUSTED VERTICAL PARAMETERS :
VERTICAL AMPLITUDE, VERTICAL POSITION AND S-CORRECTION

- PANNING (SUBTITLE VERSION) FOR VERTICAL DEFLECTION

 




- EAST-WEST FUNCTION GENERATOR WITH
INTEGRATED ERROR AMPLIFIER (THUS,
ONLY 1 EXTERNAL POWER DARLINGTON
IS NECESSARY FOR Ew FUNCTION)

- BUS CONTROLLED EAST—WEST FUNCTIONS : EW AMPLITUDE, HORIZONTAL
WIDTH, EW TILT AND Ew SHAPE (CORNER
CORRECTION)

- VAFIIABLE FIX POINTS FOR s-cORRECTION VERTICAL BLANKING AND EASTWEST CORNER CORRECTION

- SUPER SANDCASTLE OUTPUT (5V LEVEL)
TO CONTROL OTHER CIRCUITS LIKE
CHROMA DECODER,

- BUS REPLY FOR VIDEO IDENT, 50/6OHz
VERTICAL MODE, TV STANDARD

SMPS PART

I SMPS PULSE WIDTH MODULATOR WITH
BUS CONTROLLED REFERENCE VOLTAGE

I SOFT START FOR SMPS MODULATOR AND
HORIZONTAL OUTPUT STAGE

I STANDBY OPERATION OF IC

I PROTECTION CIRCUIT FOR OVERLOAD
DETECTION (SHORT-CIRCUITS) AND CONTROL OF SMPS MODULATOR

I BUS REPLY FOR OVERLOAD AND POWER
FAIL

2'19 I17 II%Ra%m R

PSI PART

I EDGE REPLACEMENT FOR Y PATH

I PEAKING FOR Y PATH WITH ADAPTIVE
CORING

I BLACK STRETCH FOR Y PATH WITH EXTERNAL TIME CONSTANT

I WHITE STRETCH FOR Y PATH WITH EXTERNAL TIME CONSTANT

I CTI (COLOR TRANSIENT IMPROVEMENT)
FOR U AND V PATH

I COMPENSATION DELAY IN Y PATH FOR
SIGNAL DELAY IN THE CHFDMA PROCESSOR

I ALL FUNCTIONS CONTROLLABLE AND
SWITCHABLE BY BUS
 
TEA2261 SWITCH MODE POWER SUPPLY CONTROLLER:icc9
The control means IP1 provide a soft start for a safe start-up after switching on the line power. This is accomplished via a resistor R5 charging slowly a capacitor C14 with a high capacitance which provides the necessary power for the integrated circuit IP1 at pins 15 and 16.

Additionally the SMPS starts with a low oscillating frequency to avoid a current build-up in the switching transistor T1. A current build-up can arise when the energy stored in the primary inductance is not fully transferred to the secondary side before a new conduction period is initiated. This will lead to operation in continuous mode and the switching transistor T1 may leave therefore his safe operating area. To reduce the oscillating frequency during start-up, the SMPS includes a resistor R511 and a diode D9 in series which connect the capacitor C26 with a capacitor C12 which is charged by the feed-back winding W2. The capacitor C12 is not charged up initially when the SMPS is switched on. Therefore, the diode D9 disconnects capacitor C26 from capacitor C12. The operating frequency is then fixed by R13 and C26, which is a low frequency (a few kHz). After a certain time capacitor C12 is charged up and then D9 will be conducting and an additional current will charge C26 via R511, thus the oscillating frequency increases to its normal operating frequency (about 22 kHz). This ensures that the SMPS starts safely in discontinuous mode, i.e. the energy stored in the primary inductance is always fully transferred to the secondary side before a new conduction period of transistor T1 is initiated.

The start-up of this known SMPS is depending on the charge-up time of capacitor C14 via resistor R5, therefore, depending on the voltage value of the AC mains input voltage. This leads to a quite long start-up time at a low mains input voltage.


The invention relates to a switch mode power supply (SMPS) comprising control means which include an oscillator for generating a pulse width modulated signal.

It is the object of the invention to provide a SMPS as previously described having a fast start-up time over a wide input voltage range. This object is accomplished with a switch mode power supply according to claim 1. The subclaims relate to preferred embodiments.

According to the invention, the switch mode power supply comprises a network which provides in case of a high input voltage a start-up with a low oscillation frequency only for the start-up time. After start-up, the oscillation frequency changes to the normal oscillating frequency. In case of a low input voltage, the network provides a start-up with essentially the normal oscillation frequency. This can be done without safety risk for the switching transistor because the operating voltages are low in this case. Even if a slight current build-up phenomenon occurs during start-up, the switching transistor stays in the safe operating area because of the low voltages. The network, therefore, includes means which change the oscillating frequency only in case of a high mains input voltage. No soft start is provided in case of a low mains input voltage. The frequency control of the oscillation frequency can be done advantageously by frequency control means including a transistor stage which change in case of a high mains input voltage the time constant of the oscillator network which determines the oscillation frequency.

In a special embodiment the network comprises a transistor used in inverse mode as a switching element. With this circuit arrangement an additional diode is not necessary. This utilizes the fact that the maximum collector base breakdown's voltage is distinctly higher than the maximum emitter base breakdown's voltage. The SMPS can be used especially for a TV receiver which works in a mains input voltage range of 90 V to 270 V, in a TV receiver the start-up time of the picture tube has to be considered additionally.

.POSITIVE AND NEGATIVE CURRENT UP TO
1.2A and – 2A
.LOW START-UP CURRENT
.DIRECT DRIVE OF THE POWER TRANSISTOR
.TWO LEVELS TRANSISTOR CURRENT LIMITATION
.DOUBLE PULSE SUPPRESSION
.SOFT-STARTING
.UNDER AND OVERVOLTAGE LOCK-OUT
.AUTOMATIC STAND-BY MODE RECOGNITION
.LARGE POWER RANGE CAPABILITY IN
STAND-BY (Burst mode)
.INTERNAL PWM SIGNAL GENERATOR


DESCRIPTION
The TEA2260/61 is a monolithic integrated circuit
for the use in primary part of an off-line switching
mode power supply.
All functions required for SMPS control under normal
operating,transient or abnormal conditions are
provided.
The capability of working according to the ”masterslave”
concept, or according to the ”primary regulation”
mode makes the TEA2260/61 very flexible
and easy to use. This is particularly true for TV
receivers where the IC provides an attractive and
low cost solution (no need of stand-by auxiliary
power supply).

GENERAL DESCRIPTION
The TEA2260/61 is an off-line switch mode power
supply controller. The synchronization functionand
the specificoperationin stand-bymodemake itwell
adapted to video applications such as TV sets,
VCRs, monitors, etc...
The TEA2260/61 can be used in two types of
architectures :
- Master/slave architecture. In this case, the
TEA2260/61 drives the power transistor according
to the pulse width modulated signals generated
by the secondary located master circuit. A
pulse transformer provides the feedback (see
Figure 1).
- Conventional architecture with linear feedback
signal (feedback sources : optocoupler or transformer
winding) (see Figure 2).
Using the TEA2260/61, the stand-by auxiliary
power supply, often realized with a small but costly
50Hz transformer, is no longer necessary. The
burst mode operation of the TEA2260/61 makes
possible the control of very low output power (down
to less than 1W) with the main power transformer.
When used in a master/slave architecture, the
TEA2260/61and also the power transistor turn-off
can be easily synchronized with the line transformer.
The switching noise cannot disturb the
picture in this case.
As an S.M.P.S.controller, the TEA2260/61features
the following functions :
- Power supply start-up (with soft-start)
- PWM generator
- Direct power transistor drive (+1.2A, -2.0A)
- Safety functions : pulse by pulse current limitation,
output power limitation, over and under voltage
lock-out.
S.M.P.S. OPERATING DESCRIPTION
Starting Mode - Stand By Mode
Power for circuit supply is taken from the mains
through a high value resistor before starting. As
long as VCC of the TEA2260/61 is below VCC start,
the quiescent current is very low (typically 0.7mA)
and the electrolytic capacitor across VCC is linearly
charged. When VCC reaches VCC start (typically
10.3V), the circuit starts, generating output pulses
with a soft-starting. Then the SMPS goes into the
stand-bymode and the output voltage is a percentage
of the nominal output voltage (eg. 80%).
For this the TEA2260/61 contains all the functions
required for primary mode regulation : a fixed frequency
oscillator, a voltage reference, an error
amplifier and a pulse width modulator (PWM).
For transmission of low power with a good efficiency
in stand-by, an automatic burst generation
system is used, in order to avoid audible noise.
Normal Mode (secondary regulation)
The normal operating of the TV set is obtained by
sending to the TEA2260/61regulation pulses generated
by a regulator located in the secondary side
of the power supply.
This architectureuses the ”Master-slave Concept”,
advantages of which are now well-known especially
the very high efficiency in stand-bymode, and
the accurate regulation in normal mode.
Stand-by mode or normal mode are obtained by
supplying or not the secondary regulator. This can
be ordonneredfor exemple by a microprocessor in
relation with the remote control unit.
Regulation pulses are applied to the TEA2260/61
through a small pulse-transformer to the IN input
(Pin 2). This input is sensitive to positive square
pulses. The typical threshold of this input is 0.85V.
The frequency of pulses coming from the secondary
regulator can be lower or higher than the
frequency of the starting oscillator.
The TEA2260/61has no soft-starting system when
it receives pulses from the secondary. The softstarting
has to be located in the secondary regulator.
Due to the principle of the primary regulation,
pulses generated by the starting system automatically
disappear when the voltage delivered by the
SMPS increases.
Stand-by Mode - Normal Mode Transition
During the transition there are simultaneously
pulses coming from the primary and secondary
regulators.
These signals are not synchronizedand some care
has to betaken toensure the safety of theswitching
power transistor.
Avery sure and simple way consist in checking the
transformer demagnetization state.
- A primary pulse is taken in account only if the
transformer is demagnetized after a conduction
of the power transistor required by the secondary
regulator.
- A secondary pulse is taken in account only if the
transformer is demagnetized after a conduction
of the power transistor required by the primary
regulator.
With this arrangement the switching safety area of
the power transistor is respected and there is no
risk of transformer magnetization.
The magnetization state of the transformer is
checked by sensing the voltage across a winding
of the transformer (generally the same which supplies
the TEA2261). This is made by connecting a
resistor between this winding and the demagnetization
sensing input of the circuit (Pin 1).


SECURITY FUNCTIONS OF THE TEA2261 (see flow-chart below)
- Undervoltage detection. This protection works
in association with the starting device ”VCC
switch” (see paragraph Starting-mode - standby
mode). If VCC is lower than VCCstop (typically
7.4V) output pulses are inhibited, in order to avoid
wrong operation of the power supply or bad
power transistor drive.
- Overvoltage detection. If VCC exceedsVCCmax
(typically 15.7V) output pulses are inhibited. Restarting
of the power supply is obtained by reducing
VCC below VCCstop.
- Current limitation of the power transistor. The
current is measured by a shunt resistor. Adouble
threshold system is used :
- When the first threshold (VIM1) is reached, the
conduction of the power transistor is stopped
until the end of the period : a new conduction
signal is needed to obtain conduction again.
- Furthermore as long as the first threshold is
reached (it means during several periods), an
external capacitor C2 is charged. When the
voltage across the capacitor reaches VC2 (typically
2.55V) the output is inhibited.This is called
the ”repetitive overload protection”. If the overload
diseappears before VC2 is reached, C2 is
discharged, so transient overloads are tolerated.
- Second current limitation threshold (VIM2).
When this thresholdis reached the output of the
circuit is immediatly inhibited. This protection is
helpfull in case of hard overload for example to
avoid the magnetization of the transformer.
- Restart of the power supply. After stopping due
to VC2, VIM2, VCCMax or VCCstop triggering, restart
of the power supply can be obtained by the
normal operating of the ”VCC switch” but thanks
to an integrted counter, if normal restart cannot
be obtained after three trials, the circuit is definitively
stopped. In this case it is necessary to
reduce VCC below approximately 5V to reset the
circuit. From a practical point of view, it means
that the power supply has to be temporarily disconnected
from any power source to get the
restart.



NORDMENDE (THOMSON) CONTURA 63 (ICC9)  CHASSIS ICC9  Television startup current regulation:
 A television receiver has a switched mode power supply controller, which may be deflection synchronized, which produces pulses in a power transformer. To power the controller during startup, before the internal power supply has started, a storage capacitor on the power supply input of the controller is coupled through a current limiting resistance to a rectified voltage from the AC mains so as to charge the storage capacitor during first polarity phases of the AC mains. The current limiting resistance includes a positive temperature coefficient (PTC) element which increases its impedance as it heats, to reduce power dissipation after initial connection of the television receiver to AC mains. In addition, a diode is coupled in a current path to the storage capacitor to charge the capacitor to the same polarity as during opposite polarity phases of the AC mains. The two paths for charging the capacitor allow a relatively smaller value for the current limiting resistance, which remains coupled to the storage capacitor, and provides sufficient power to the controller without undue power dissipation over a wide range of power mains voltages.

1. A television apparatus having a degaussing coil for demagnetizing metal portions of a cathode ray tube, said degaussing coil being energized from an AC source, and coupled to a first temperature dependent impedance for controlling the current passing in said degaussing coil, said first temperature dependent impedance being thermally coupled to a second temperature dependent impedance for heating said first temperature dependent impedance, a power supply including a capacitance for energizing a load circuit, and means for charging said capacitance through said second temperature dependent impedance, said capacitor being additionally charged through a path including a rectifier, said path being independent of said first and second temperature dependent impedances.

Description:
The invention relates to the startup of a switched mode power supply in a television receiver.
In a television receiver having a switched mode power supply (SMPS), which may be deflection synchronized, the primary winding of a power transformer is energized by a pulse width modulated signal. The secondary windings of the transformer energize DC power supplies which provide power to operational loads, including the SMPS controller. In the start up interval, immediately after switching the television receiver on, it is necessary to initiate generation of power through the power transformer in order to begin operation. The SMPS controller itself may be powered from a storage capacitor which is initially charged by rectified AC mains current through a current limiting resistor. The capacitor charges initially when the television receiver is connected to the AC mains (i.e., plugged in) and remains charged for energizing the controller whenever the television receiver is either in the standby mode or run mode of operation.

The current limiting resistor dissipates power as long as the television receiver is coupled to the AC mains. However, for purposes other than providing power during the startup interval, this power is wasted. The current limiting resistor can have a high resistance to reduce power dissipation, but a higher resistance results in a reduced current supply for operation of the controller, and slower charging of the storage capacitor. It is necessary to reconcile the need for current to the controller in the startup interval with the need to reduce power dissipation in the current limiting resistor at all other times.
In designing a switched mode power supply, it is advantageous to provide a single circuit that is operable over a range of different mains voltages. The standard mains voltages for the US and for Europe, for example, differ substantially. A circuit which is optimal at one mains voltage may include current limiting elements which produce excessive power dissipation, inadequate current supply or other adverse effects when operated at a different mains voltage.

Apart from circuitry associated with startup of operational power, television receivers are typically provided with degaussing coils which demagnetize ferromagnetic parts of the picture tube to improve color purity. The degaussing coils may be coupled to the AC mains through one or more variable impedance elements that progressively reduce the current applied to the degaussing coils over a degaussing interval following the connection of the television to the AC mains. The current limiting elements can be positive or negative temperature coefficient resistors, also known as thermistors. In one technique, a first temperature dependent resistor having a positive coefficient is coupled to the AC mains in series with the degaussing coil. A second temperature dependent resistor having a positive coefficient is coupled across the AC mains, the two temperature dependent resistors being thermally coupled to each other such that each heats the other. As the resistances of the elements change with heating, current through the degaussing coil falls off to a minimum level which does not substantially affect color purity.
In an inventive arrangement, temperature dependent elements may be used in connection with the charging of a storage capacitor for the supply voltage of a switched mode power supply controller, as a means to limit power dissipation. In carrying out an inventive feature, operation of a television receiver during startup is optimized by coupling the degaussing circuitry with the current limiting means for charging the storage capacitor that provides power to the switched mode power supply controller. A circuit for degaussing and current limited startup supply, optimized over a wide range of power mains voltages may thus be achieved.
In accordance with an aspect of the invention, a television apparatus has a degaussing coil for demagnetizing metal portions of a cathode ray tube. The degaussing coil is energized from an AC source, and coupled to a first temperature dependent impedance for controlling the current passing in the degaussing coil. The first temperature dependent impedance is thermally coupled to a second temperature dependent impedance for heating the first temperature dependent impedance. A power supply includes a capacitance for energizing a load circuit. The capacitance is charged through the second temperature dependent impedance.
In accordance with another aspect of the invention, a television apparatus having a power supply includes a capacitance for energizing a load circuit and a source for charging the capacitance. A start-up circuit comprises a source of AC current. A first polarity of the AC current is passed to charge the capacitance, and an opposite polarity of the AC current is passed through a temperature sensitive impedance via a path independent of the path of the first polarity of the AC current, to charge the capacitor in the same direction as said first polarity of said AC current.

FIG. 1 is a schematic circuit diagram of part of a television receiver incorporating a startup current supply according to the invention;
FIG. 2 is a graph of voltage vs. time in the circuit shown in FIG. 1 at the junction of current limiting resistor R2 and diode D7 with respect to ground; and
FIG. 3 is a graph of voltage vs. time at the same point as in FIG. 2, but at a higher mains voltage.
In a television receiver as shown in FIG. 1, power for the operational loads is derived from a power transformer X1 when driven by a switched mode power supply controller 60 and a power output transistor Q1. In some applications, the power transformer may be a horizontal output transformer. The various operational loads are coupled to secondary windings of transformer X1, one operational load RL being shown as coupled to a secondary winding 34. A diode D5 and a filter capacitor C3 are coupled to the secondary winding 34 for supply of regulated DC voltage to load RL. Additional secondary windings typically are provided for power supply at different voltage levels, as required for operating the loads. Only two windings, 34 and 36, are shown in order to simplify the drawing.
The supply voltage Vin to transformer X1 is derived from a full wave bridge rectifier formed by diodes D1 through D4, coupled to the AC mains 22 through plug 23, surge suppressor chokes L1, L2, and bypass capacitor C1. The full wave rectified output, at the cathodes of diodes D2 and D4 of the bridge rectifier, is coupled to the primary winding 32 of power transformer X1 through current limiting resistor R1 and filter capacitor C2. Supply voltage Vin is available whenever AC mains 22 is connected to the television receiver, but power to the loads RL is provided only after switched mode controller 60 becomes energized.
Secondary winding 36 of transformer X1 energizes the switched mode power supply controller 60 after initial startup, via diode D6. However, this voltage is available only during switching of transistor Q1. Since the supply of power to the controller is arranged functionally in a loop where output pulses of the controller are required before power can be provided through secondary winding 34, at startup, an alternate source of power to controller 60 is required.
The television receiver includes a degaussing circuit 21 which is energized for a brief interval following connection of AC mains 22 to the television receiver, for demagnetizing the ferromagnetic elements of a picture tube 40. The degaussing circuit includes a degaussing coil L3 coupled to AC mains 22 and variable resistance elements V1, V2, operable to reduce current to the degaussing coil over time such that AC current supplied to coil L3 starts at a high amplitude and then falls off to a minimum. The current limiting elements V1, V2 are positive temperature coefficient (PTC) resistors or thermistors, and are mounted in thermal contact with one another, as shown by broken line 50, such that the heat generated by each contributes to increasing the resistance of both.
One of the current limiting elements, V2, is coupled in series with the degaussing coil, the series branch being coupled across AC mains 22. When the television receiver is first connected to mains 22, the resistance of element V2 is low, and increases with heating due to dissipation of power with current flow through degaussing coil L3. The other variable resistance element, V1, is coupled in series with diode D7, the series branch being in parallel with AC mains 22, and adds to the current passing through element V1, and therefore adds to the heating of element V2. Diode D7 blocks current through element V1 during each alternate phase of the voltage supplied by AC mains 22.
In carrying out an aspect of the invention, the cathode of diode D7 is coupled to a current limiting resistor R2 at a terminal 17 to provide startup current to SMPS controller 60. Diode D7, in cooperation with diode D3 of the mains bridge, forms a half wave rectifier of the AC mains voltage, that in conjunction with resistor R2 provides a first path for low level DC current to charge filter capacitor C5. Diode D7 and D3 conduct during the positive phase of the AC mains voltage, i.e. when terminal 15 is positive relative to terminal 16.
During the alternate negative phase of the AC mains voltage, when terminal 15 is negative relative to terminal 16, diode D1, of the mains bridge, provides half wave rectification of the AC mains voltage to resistor R2 via PTC resistor V1. Diodes D1 and PTC resistor V1 form a second current path which provides a low level DC current to charge filter capacitor C5.
By means of the two alternating conducting current paths, capacitor C5 charges to a level that provides adequate operating voltage to SMPS controller 60, to begin free running power supply operation.
Rectifying diode D6, coupled to secondary winding 36 of transformer X1, blocks discharge of capacitor C5 through winding 36 when SMPS controller 60 is not energizing transformer X1, and provides the main charging path for capacitor C5 when the SMPS controller is energizing the transformer.
The circuit shown is advantageous because it operates over a wide range of mains voltages and is effective at both high mains voltage and at low mains voltage for providing adequate current to charge storage capacitor C5, while reducing unnecessary power dissipation in current limiting resistor R2. This aspect of the invention may be appreciated by comparing the curves of FIGS. 2 and 3, which show the voltages over time at the cathode of diode D7 at a lower mains voltage, e.g., about 90 V RMS (FIG. 2) and at a higher mains voltage, e.g., about 270 V RMS (FIG. 3). The curves represent the steady state operation of the circuit, after PTC resistors V1, V2 have reached their maximum temperature and resistance values.
In both FIGS. 2 and 3, the voltage at the cathode of diode D7 is higher in the positive phase 84, 94 of power on AC mains 22 than in the negative phase 82, 92. This occurs because in the positive phase, diode D7 is forward biased and the voltage applied to resistor R2 at terminal 17 is equal to the mains voltage. In the negative phase, PTC resistor V1 is coupled between the mains and terminal 17 and absorbs some of the mains voltage before it is applied to resistor R2.
At the relatively lower mains voltage in FIG. 2, the PTC resistors are heated to a relatively lower temperature than at the higher mains voltage of FIG. 3 because the extent of heating is a function of the power dissipation in the PTC resistors. Since the heating of the PTC resistors is a nonlinear effect, the substantially higher resistance of PTC resistor V1 at the higher mains voltage and temperature is such that the voltage applied to resistor 12 in the negative phase is proportionately much lower than the voltage in the positive phase when operating at the higher mains voltage. Thus, the relative amplitude of 82 to 84 at the lower mains voltage is closer to unity than the relative amplitude of 92 to 94 at the higher mains voltage. The difference in the illustrated example is such that power dissipated in current limiting resistor R2 is reduced by about 25% over the range of mains voltages from 90 VAC to 250 VAC.
It should be noted that the startup circuit of the instant invention uses the PTC resistor V1, which is part of the degaussing circuitry. As a result, the startup circuit requires only a few parts in addition to the degaussing circuitry.
It should be further noted that diode D7 reduces the power dissipated by PTC resistor V1 by about half. Nevertheless, the degaussing function is not impaired. The residual current through degaussing coil L3 after the end of the degaussing interval is insignificant.
In some applications, it may be desirable to operate the SMPS in synchronism with deflection in order to prevent switching transients from appearing on the display screen. In such an arrangement, the SMPS would free-run during start-up until stable synchronization signals become available.
 


NORDMENDE (THOMSON) CONTURA 63 (ICC9)  CHASSIS ICC9  Synchronized switch-mode power supply:

In a switch mode power supply, a first switching transistor is coupled to a primary winding of an isolation transformer. A second switching transistor periodically applies a low impedance across a second winding of the transformer that is coupled to an oscillator for synchronizing the oscillator to the horizontal frequency. A third winding of the transformer is coupled via a switching diode to a capacitor of a control circuit for developing a DC control voltage in the capacitor that varies in accordance with a supply voltage B+. The control voltage is applied via the transformer to a pulse width modulator that is responsive to the oscillator output signal for producing a pulse-width modulated control signal. The control signal is applied to a mains coupled chopper transistor for generating and regulating the supply voltage B+ in accordance with the pulse width modulation of the control signal.

Description:

The invention relates to switch-mode power supplies.

Some television receivers have signal terminals for receiving, for example, external video input signals such as R, G and B input signals, that are to be developed relative to the common conductor of the receiver. Such signal terminals and the receiver common conductor may be coupled to corresponding signal terminals and common conductors of external devices, such as, for example, a VCR or a teletext decoder.

To simplify the coupling of signals between the external devices and the television receiver, the common conductors of the receiver and of the external devices are connected together so that all are at the same potential. The signal lines of each external device are coupled to the corresponding signal terminals of the receiver. In such an arrangement, the common conductor of each device, such as of the television receiver, may be held "floating", or conductively isolated, relative to the corresponding AC mains supply source that energizes the device. When the common conductor is held floating, a user touching a terminal that is at the potential of the common conductor will not suffer an electrical shock.

Therefore, it may be desirable to isolate the common conductor, or ground, of, for example, the television receiver from the potentials of the terminals of the AC mains supply source that provide power to the television receiver. Such isolation is typically achieved by a transformer. The isolated common conductor is sometimes referred to as a "cold" ground conductor.

In a typical switch mode power supply (SMPS) of a television receiver the AC mains supply voltage is coupled, for example, directly, and without using transformer coupling, to a bridge rectifier. An unregulated direct current (DC) input supply voltage is produced that is, for example, referenced to a common conductor, referred to as "hot" ground, and that is conductively isolated from the cold ground conductor.
A pulse width modulator controls the duty cycle of a chopper transistor switch that applies the unregulated supply voltage across a primary winding of an isolating flyback transformer. A flyback voltage at a frequency that is determined by the modulator is developed at a secondary winding of the transformer and is rectified to produce a DC output supply voltage such as a voltage B+ that energizes a horizontal deflection circuit of the television receiver. The primary winding of the flyback transformer is, for example, conductively coupled to the hot ground conductor. The secondary winding of the flyback transformer and voltage B+ may be conductively isolated from the hot ground conductor by the hot-cold barrier formed by the transformer.

It may be desirable to synchronize the operation of the chopper transistor to horizontal scanning frequency for preventing the occurrence of an objectionable visual pattern in an image displayed in a display of the television receiver.

It may be further desirable to couple a horizontal synchronizing signal that is referenced to the cold ground to the pulse-width modulator that is referenced to the hot ground such that isolation is maintained.

A synchronized switch mode power supply, embodying an aspect of the invention, includes a transfromer having first and second windings. A first switching arrangement is coupled to the first winding for generating a first switching current in the first winding to periodically energize the second winding. A source of a synchronizing input signal at a frequency that is related to a deflection frequency is provided. A second switching arrangement responsive to the input signal and coupled to the second winding periodically applies a low impedance across the energized second winding that by transformer action produces a substantial increase in the first switching current. A periodic first control signal is generated. The increase in the first switching current is sensed to synchronize the first control signal to the input signal. An output supply voltage is generated from an input supply voltage in accordance with the first control signal.


PHILIPS TDA4671 Picture Signal Improvement (PSI) circuit:

 GENERAL DESCRIPTION
The TDA4671 delays the luminance signal and improves
colour-difference signal transients. The luminance signal
can also be improved by peaking and noise reduction
(coring).


FEATURES
• Luminance signal delay from 20 to 1100 ns (minimum
step 45 ns)
• Luminance signal peaking with symmetrical overshoots
selectable
• Selectable 2.6 or 5 MHz peaking centre frequency and
degree of peaking (−3, 0, +3 and +6 dB)
• Selectable noise reduction by coring
• Handles negative as well as positive colour-difference
signals
• Selectable Colour Transient Improvement (CTI) to
decrease the colour-difference signal transient times to
those of the high frequency luminance signals
• Selectable 5 or 12 V sandcastle input voltage
• All controls selected via the I2C-bus
• Timing pulse generation for clamping and delay time
control synchronized by sandcastle pulse
• Automatic luminance signal delay correction using a
control loop
• Luminance and colour-difference input signal clamping
with coupling capacitor
• 4.5 to 8.8 V supply voltage range
• Minimum of external components.

FUNCTIONAL DESCRIPTION
The TDA4671 contains luminance signal processing and
colour-difference signal processing. The luminance signal
section comprises a variable, integrated luminance delay
line with luminance signal peaking and a noise reduction
by coring.
The colour-difference section consists of a transient
improvement circuit to decrease the rise and fall times of
the colour-difference signal transients. All functions and
parameters are controlled via the I2C-bus.
Y-signal path
The video and blanking signal is AC-coupled to the input
pin 16. Its black porch is clamped to a DC reference
voltage to ensure the correct operating range of the
luminance delay stage.
The luminance delay line consists of all-pass filter sections
with delay times of 45, 90, 100, 180 and 450 ns
(see Fig.1). The luminance signal delay is controlled via
the I2C-bus in steps of 45 ns in the range of 20 to 1100 ns,
this ensures that the maximum delay difference between
the luminance and colour-difference signals is ±22.5 ns.
An automatic luminance delay time adjustment in an
internal control loop (with the horizontal frequency as a
reference) is used to correct changes in the delay time,
due to component tolerances. The control loop is
automatically enabled between the burst key pulses of
lines 16 (330) and 17 (331) during the vertical blanking
interval. The control voltage is stored in capacitor CDL
connected to pin 2.
The peaking section is using a transversal filter circuit with
selectable centre frequencies of 2.6 and 5.0 MHz.
It provides selectable degrees of peaking of −3, 0, +3 and
+6 dB and noise reduction by coring, which attenuates the
high-frequency noise introduced by peaking.
The output buffer stage ensures a low-ohmic VBS output
signal on pin 12 (<160 br="" gain="" luminance="" of="" the="">signal path from pin 16 to pin 12 is unity.
An oscillation signal of the delay time control loop is
present on output pin 12 instead of the VBS signal. It is
present during the vertical blanking interval of the burst key
pulses in lines 16 (330) to 18 (332). This sync should not
be applied for synchronization.
Colour-difference signal paths
The colour-difference input signals (on pins 3 and 7) are
clamped to a reference voltage.
Each colour-difference signal is fed to a transient detector
and to an analog signal switch with an attached voltage
storage stage.
The transient detectors consist of differentiators and
full-wave rectifiers. The output voltages of both transient
detectors are added and then compared. The comparator
controls both following analog signal switches
simultaneously.
The analog signal switches are in open position at a
certain value of transient time; the held value (held by
capacitors) is then applied to the outputs. The switches
close to rapidly accept the actual signal levels at the end
of these transients. The improved transient time is
approximately 100 ns long independent of the input
transient time.
Colour-difference paths are independent of the input
signal polarity and have a nominal unity gain.
The CTI functions are switched on and off via the I2C-bus.
 In a method of picture signal improvement, an input picture signal (Yin) is filtered (lpf, S) to obtain a high-frequency component (Yin-Ylpf) and a low-frequency component (Ylpf), a standard deviation signal (SD) is calculated from the input picture signal (Yin), the high-frequency component (Yin-Ylpf) is gain controlled (MS, MP) in dependence upon the standard deviation signal (SD) to obtain a gain-controlled high-frequency component (Gcntr*Ghpf * (Yin-Ylpf)), and the gain-controlled high-frequency component (Gcntr*Ghpf * (Yin-Ylpf)) and the low-frequency component (Ylpf) are combined (A) to obtain an output signal (Yout).

1. A method of picture signal improvement, the method comprising the steps of: filtering (lpf, S) an input picture signal (Yin) to obtain a high-frequency component (Yin-Ylpf) and a low-frequency component (Ylpf); calculating a standard deviation signal (SD) from the input picture signal (Yin); gain controlling (MS, MP) the high-frequency component (Yin-Ylpf) in dependence upon the standard deviation signal (SD) to obtain a gain-controlled high-frequency component (Gcntr*Ghpf * (Yin-Ylpf)); and combining (A) the gain-controlled high-frequency component (Gcntr*Ghpf * (Yin-Ylpf)) and the low-frequency component (Ylpf) to obtain an output signal (Yout).

2. A method as claimed in claim 1, wherein the gain controlling step (MS, MP) comprises the step of gain controlling (MS, MP) the high-frequency component (Yin-Ylpf) in dependence upon the standard deviation signal (SD) and an amount of noise to obtain the gain-controlled high-frequency component (Gcntr*Ghpf * (Yin-Ylpf)).

3. A device for picture signal improvement, the device comprising: means for filtering (lpf, S) an input picture signal (Yin) to obtain a high-frequency component (Yin-Ylpf) and a low-frequency component (Ylpf); means for calculating a standard deviation signal (SD) from the input picture signal (Yin); means for gain controlling (MS, MP) the high-frequency component (Yin-Ylpf) in dependence upon the standard deviation signal (SD) to obtain a gain-controlled high-frequency component (Gcntr*Ghpf * (Yin-Ylpf)); and means for combining (A) the gain-controlled high-frequency component (Gcntr*Ghpf * (Yin-Ylpf)) and the low-frequency component (Ylpf) to obtain an output signal (Yout).

4. A still picture camera, comprising: an image pick-up device (PU) to obtain an input picture signal (Yin); and a device for improving the input picture signal (Yin) as claimed in claim 3.

Description:
SU-A-1099418 discloses a TV gradation distortions corrector that has outputs from differentiating circuits thresholded for coincidence gate controlling switching of signal free from noise components. The input TV signal is fed to the delay line and low-pass filters. The h.f. component signal passes to the commentator, and the l.f. component signal to the additional adder. There is no need for non-linear processing of signals from details for which the peak-to-peak amplitude of the video signals exceeds three times the standard deviation of the noise level. The commutator is controlled by the thresholds and coincidence gate so that the components are eliminated. Use/advantage: in communications engineering, to increase the definition and sharpness of details of the dark parts of an image, so increasing accuracy of correction.
EP-A-0,560,170 discloses a method for sharpness enhancement, whereby correction values derived from the difference between spatially neighbored pixel values are combined with the pixel values, in which the correction values are calculated according to a non-linear function of the difference values, and the pixel values are modified only if at minimum one of the surrounding pixels is correlated with the actual pixel, whereby the correction values are limited to a maximum correction value, and the correction value is added to and subtracted from, respectively, the neighboring pixels in a symmetrical way to improve the appearing sharpness of the resulting transition.
It is, inter alia, an object of the invention to provide a sharpness increase in such a manner that noise in reduced. To this end, the invention provides a picture improvement as defined in the independent claims. Advantageous embodiments are defined in the dependent claims.
In a method of picture signal improvement in accordance with a first aspect of the invention, an input picture signal is filtered to obtain a high-frequency component and a low-frequency component, a standard deviation signal is calculated from the input picture signal, the high-frequency component is gain controlled in dependence upon the standard deviation signal to obtain a gain-controlled high-frequency component, and the gain-controlled high-frequency component and the low-frequency component are combined to obtain an output signal.

THOMSON STV2151 FULL AUTOMATIC MULTISTANDARD CHROMA
DECODER WITH EMBEDDED CHROMA DELAY LINE,


 DESCRIPTION
The STV2151 integrates in a single chip every
circuitry to deliver the Y, R-Y, B-Y signals starting
trom a CVBS or Y/C signals.



It can process PAL, SECAM and NTSC standards.

It is controlled by I2C Bus.

- Inputs: one input is dedicated to the CVBS or Y
signal. An other one inputs a C signal. An inte-
grated switch, controlled by BUS, allowsto chose
the right input. According to the application, this
operation can be automatically treated by the
microprocessor, thanks a standard identification
reply available in a FC Bus register.

The synchronisation is done through a Super
Sand Castle input.

- Luminance Path : depending on the current
decoding stand ard, a colour sub-carrier trap
(notch filter), totally integrated and alignmentfree,
can be used or by-passed (BUS control)to deliver
the Y output signal.

- Chroma decoder : the chroma signal goes first
through the band pass filter ("belt filter” for SE-
CAM), which is automatically tuned by the
STV2151. It is then directly fit into the muItistan-
dard decoder. At least. the demodulated signals
are delayed in the integrated base band delay line
or led into an adder to deliver the R-Y and B-Y
signals.

In NTSC, the hue control allows a typical phase
shift of 130°.

FULL AUTOMATIC MULTISTANDARD CHROMA
DECODER WITH EMBEDDED CHROMA DELAY LINE

 COLOR DECODER FOR STANDARDS:

- SECAM

- PAL B,G

— NTSC 3.58

- PAL M

- NTSC 4.43

I TWO MODES OF SELECTION OF THE
STANDARDS, SELECTED BY BUS :

- Automatic sequentialselection mode on
SECAM/PAL B, G with NTSC 3.58 selected
by 60Hz bit only

- BUS IC standard selection mode for:
SECAM/PAL B, GI NT SC 3.58 I PAL M /
NTSC 4.43

I AUTOMATIC STANDARD RECOGNITION

I INTEGRATED CHROMA DELAY LINES IN BASE BAND

I COLOR SUB—CARRIER REGENERATION WITH XT AL (4.43 and 3.58)

I AGC FOR SECAM

I HUE CONTROL 1 aodeg FOR NTSC

I S-VHS INPUT (Bus Selection)

I AUTO ALIGNED CHROMA FILTERS

I INTEGRATED AND ADJUSTMENT FREE

TRAP FILTERS
I BIDIRECTIONAL BUS INFORMATION :
- Input Data: Standard bits
50/60Hz Bit
Auto Mode for Standard
Forced Killer Mode
Killer On/Off
Bell Filter
Central Frequency
Hue Control Bits
S-VHS Mode
- Output Data: Selected Standard Bits



PHILIPS TDA9820 Multistandard/dual channel TV FM intercarrier sound demodulator,

GENERAL DESCRIPTION
The TDA9820 is a monolithic, integrated, multistandard TV
FM intercarrier sound demodulator for all FM standards.
The circuit contains two separate FM demodulators using
Phase Locked Loop (PLL) reference frequency
generation. The circuit requires a minimum number of
external components.


FEATURES
• Multistandard application for sound standards M, B/G, I
and D/K
• Two alignment-free PLL FM demodulators
• Four-input source selector for one of the two FM
demodulators
• Automatic second sound carrier mute
• Mono and dual channel application
• Low power consumption
• Few external components required.

FUNCTIONAL DESCRIPTION
The complete circuit consists of two separate channels,
each consisting of a limiter-amplifier, FM demodulator and
AF amplifier. Circuit operation is also described in Fig.1.
Source selector
The intercarrier signal is fed through external ceramic
band-pass filters which are tuned to the sound carrier
frequencies.
One of the four filtered sound carriers from
pins 1, 2, 3 or 16 is fed to limiter-amplifier 1 via the
appropriate electronic switch in the source selector.
The electronic switch of the sound carrier is selected by
the control unit (see Table 1).
The second sound carrier of the intercarrier signal is
directly fed from pin 15 to limiter-amplifier 2.
FM demodulators
Each limiter-amplifier is AC-coupled into a
FM demodulator. The integrated FM demodulator PLLs
are alignment-free. The FM demodulator outputs are
amplified to 500 mV (RMS value). High amplification and
DC error signals of the PLLs, which are superimposed on
the FM demodulator outputs, require DC decoupling at
pins 9 and 10 of the AF amplifier inputs.
Stereo channel separation adjustment (optional)
Optimal stereo channel separation is achieved by
adjusting VAF1 (pin 8) and VAF2 (pin 7) as follows:
1.
VAF1 by a resistor in series with the DC decoupling
capacitor at pin 9
2.
VAF2 by a variable resistor in series with the
DC decoupling capacitor on pin 10 to the same
voltage as VAF1.
Second sound carrier mute
The output of the second FM demodulator is muted when
the signal level (signal and/or noise) at pin 15 is less than
typically 0.5 mV (RMS value). This avoids an incorrect
stereo or dual sound identification when a mono signal is
transmitted. Therefore, with a mono transmission, there is
no audio output at pin 7. When the signal level at pin 15 is
greater than typically 1.0 mV (RMS value) mute is
switched off.
Control unit
The control unit selects the required sound standard
according to the voltages on pin 5 and pin 6. The control
unit performs the following:
1.
selects the free-running frequencies of VCO1 and
VCO2
2.
switches the source selector .

PHILIPS TDA2616/TDA2616Q 2 x 12 W hi-fi audio power amplifiers with mute,

 GENERAL DESCRIPTION
The TDA2616 and TDA2616Q are dual power amplifiers.
The TDA2616 is supplied in a 9-lead single-in-line (SIL9)
plastic power package (SOT131), while the TDA2616Q is
supplied in a 9-lead SIL-bent-to-DIL plastic power package
(SOT157). They have been especially designed for mains
fed applications, such as stereo radio and stereo TV.

 FEATURES
• Requires very few external components
• No switch-on/switch-off clicks
• Input mute during switch-on and switch-off
• Low offset voltage between output and ground
• Excellent gain balance of both amplifiers
• Hi-fi in accordance with IEC 268 and DIN 45500
• Short-circuit proof and thermal protected
• Mute possibility.

FUNCTIONAL DESCRIPTION
The TDA2616 is a hi-fi stereo amplifier designed for mains
fed applications, such as stereo radio and TV. The circuit
is optimally designed for symmetrical power supplies, but
is also well-suited to asymmetrical power supply systems.
An output power of 2 × 12 W (THD = 0.5%) can be
delivered into an 8 Ωload with a symmetrical power supply
of ±16 V. The gain is internally fixed at 30 dB, thus offering
a low gain spread and a very good gain balance between
the two amplifiers (0.2 dB).
A special feature is the input mute circuit. This circuit
disconnects the non-inverting inputs when the supply
voltage drops below ±6 V, while the amplifier still retains its
DC operating adjustment. The circuit features suppression
of unwanted signals at the inputs, during switch-on and
switch-off.
The mute circuit can also be activated via pin 2. When a
current of 300 µA is present at pin 2, the circuit is in the
mute condition.
The device is provided with two thermal protection circuits.
One circuit measures the average temperature of the
crystal and the other measures the momentary
temperature of the power transistors. These control
circuits attack at temperatures in excess of +150 °C, so a
crystal operating temperature of max. +150 °C can be
used without extra distortion.
With the derating value of 2.5 K/W, the heatsink can be
calculated as follows:
at RL = 8 Ω and VP = ±16 V, the measured maximum
dissipation is 14.6 W.
With a maximum ambient temperature of +65 °C, the
thermal resistance of the heatsink is:
The internal metal block has the same potential as pin 5.

THOMSON TDA8172 TV VERTICAL DEFLECTION OUTPUT CIRCUIT:

DESCRIPTION
The TDA8172 is a monolithic integrated circuit in
HEPTAWATTTMpackage. It is a high efficiency
power boosterfordirectdriving of verticalwindings
of TV yokes. It is intendedfor use in Color andB &
W television as well as in monitorsand displays.











TEA5101A - RGB HIGH VOLTAGE AMPLIFIER BASIC OPERATION AND APPLICATIONS:

GENERAL
The control of state-of-the-art color cathode ray
tubes requires high performance video amplifiers
which must satisfy both tube and video processor
characteristics.
When considering tube characteristics (see Fig-
ures 13 and 14),we note that a 130V cutoff voltage
is necessary to ensure a 5mA peak current.How-
ever 150V is a more appropriate value if the satu-
ration effect of the amplifier is to be taken into
account. As the dispersion range of the three guns
is ± 12%, the cutoff voltage should be adjustable
from 130V to 170V. The G2 voltage, from 700 to
1500V allows overall adjustment of the cutoff volt-
age for similar tube types.
A 200V supply voltage of the video amplifier is
necessary to achieve a correct blanking operation.
In addition, the video amplifier should have an
output saturation voltage drop lower than 15V, as
a drive voltage of 130V (resp. 115V) is necessary
to obtain a beam current of 4 mA for a gun which
has a cutoff point of 170V (resp. 130V).
Note : For all the calculations discussed above, the
G1 voltage is assumed to be 0V.
The video processor characteristics must also be
considered. As it generally delivers an output volt-
age of 2 to 3V, the video amplifier must provide a
closed loop DC gain of approximately 40.
The video amplifier dynamic performances must
also meet the requirements of good definition even
with RGB input signals (teletext,home computer...),
e.g. 1mm resolution on a 54cm CRT width scanned
in 52µs. Consequently, a slew rate better than
2000V/µs, i.e. rise and fall times lower than 50ns,
is needed. In addition, transition times must be the
same for the three channels so as to avoid coloured
transitions when displaying white characters. The
bandwidth of a video amplifier satisfying all these
requirements must be at least 7MHz for high level
signals and 10MHz for small signals.
One major feature of a video amplifier is its capa-
bility to monitor the beam current of the tube. This
function is necessary with modern video proces-
sors:
- for automatic adjustment of cutoff and also, where
required,video gain in order to improve the long
term performances by compensation for aging
effects through the life of the CRT. This adjust-
ment can be done either sequentially (gun after
gun) or in a parallel mode.
- for limiting the average beam current
A video amplifier must also be flashover protected
and provide high crosstalk performances. Cros-
stalk effects are mainly caused by parasitic capaci-
tors and thus increase with the signal frequency. A
crosstalk level of -20dB at 5MHz is generally ac-
ceptable.

Table 1 summarizes the main features of a high
performance video amplifier.
Table 1 :
Main Features of a High Performance
Video Amplifier
Maximum Supply Voltage
220V
Output voltage swing "Average"
100V
Output voltage swing "Peak"
130V
Low level saturation (refered to VG1)
15V
Closed loop gain
40
Transition time
50ns
Large signal bandwidth
7MHz
Small signal bandwidth
10MHz
Beam current monitoring
Flash over protection
Crosstalk at 5MHz
-20dB
The SGS-THOMSON Microelectronics TEA5101A
is a high performance and large bandwidth 3 chan-
nel video amplifier which fulfills all the criteria dis-
cussed above. Designed in a 250V DMOS bipolar
technology, it operates with a 200V power supply
and can deliver 100V peak-to-peak output signals
with rise and fall times equal to 50ns.
The 5101A features a large signal bandwidth of
8MHz, which can be extended to 10MHz for small
signals (50 Vpp).
Each channel incorporates a PMOS transistor to
monitor the beam current. The circuit provides
internal protection against electrostatic discharges
and high voltage CRT discharges.
The best utilization of the TEA 5101A high perform-
ance features such as dynamic characteristics,
crosstalk,or flashover protection requires opti-
mized application implementation. This aspect will
be discussed in the fourth part of this document.

I.1 - Input Stage
The differential input stage consists of the transistor
T1 and T2 and the resistors R4,R5 and R6.
This stage is biased by a voltage source T3,R1,R2
and R3.
VB(T1) = (1 + R2
R3) x VB(T3) ≅ 3.8V
Each amplifier is biased by a separate voltage
source in order to reduce internal crosstalk. The
load of the input stage is composed of the transistor
T4 (cascode configuration) and the resistor R7. The
cascode configuration has been chosen so as to
reduce the Miller input capacitance. The voltage
gain of the input stage is fixed by R7 and the emitter
degeneration resistors R5,R6,and the T1,T2 internal
emitter resistances. The voltage gain is approxi-
mately 50dB.
Using a bipolar transistor T4 and a polysilicon re-
sistor R7 gives rise to a very low parasitic capaci-
tance at the output of this stage (about 1.5pF).
Hence the rise and fall times are about 50ns for a
100V peak-to-peak signal (between 50V and
150V).
I.2 - Output Stage
The output stage is a quasi-complementary class
B push-pull stage. This design ensures a symetrical
load of the first stage for both rising and falling
signals. The positive output stage is made of the
DMOS transistor T5,and the negative output stage
is made of the transistors PMOS T6 and DMOS T7.
The compound configuration T6-T7 is equivalent to
a single PMOS. A single PMOS transistor capable
of sinking the total current would have been too
large.
By virtue of the symetrical drive properties of the
output stage the rise and fall times are equal (50ns
for 100V DC output voltage).

 I.3 - Beam Current Monitoring
This function is performed by the PMOS transistor
T8 in source follower configuration. The voltage on
the source (cathode output) follows the gate volt-
age (feedback output). The beam current is ab-
sorbed via T8 . On the drain of T8, this current will
be monitored by the videoprocessor.
I.4 - Protection Circuits
I.4.1 - MOS protection
Four zener diodes DZ(1-4) are connected between
gate and source of each MOS in order to prevent
the voltage from reaching the breakdown volt-
age.Hence the VGS voltage is internally limited to
± 15V.
I.4.2 - Protection against electrostatic dis-
charges
All the input/output pins of the TEA5101A are pro-
tected by the diodes D1-D7 which limit the overvol-
tage due to ESD.
I.4.3 - Flashover Protection
A high voltage and high current diode D5 is con-
nected between each output and the high voltage
power supply. During a flash, most of the current is
generally absorbed by the spark gap connected to
the CRT socket. The remaining current is absorbed
by the high voltage decoupling capacitor through
the diode D5. Hence the cathode voltage is
clamped to the supply voltage and the output volt-
age does not exceed this value.

 I.1 - Voltage Amplifier
II.1.1 - Bias conditions Vin = Vref
The bias point is fixed by the feedback resistor
Rf,the bias resistor Rp, and by the internal refer-
ence voltage when Vin = Vref.
If VO is the output voltage (pin 9) :
VO = (1 + Rf
Rp) x Vref (1)
In this state T1 and T2 are conducting. A current
flows in R7 and T4 soT5 is on. The T5 drain current
is fed to the amplifier input through the feedback
resistor. The current in R7 is:
I(R7) = VDD − VO − VGS(T5)
R7
≅ VDD − VO
R7
and the current in T5 and Rf is :
I(T5) = VO − Vref
Rf
≅ VO
Rf
Thus the total current absorbed by each channel of
the TEA5101A is :
VDD
R7 + VO x (1
Rf − 1
R7)
The cathode (pin 7) output voltage is:
VO + VGS(T8) = VO
The beam current is absorbed by T8 and Rm. The
voltage developed across Rm by this current is fed
to the videoprocessor in order to monitor the beam
current.
II.1.2 - Dynamic operation
The TEA5101A operates as a closed loop amplifier,
with its voltage gain fixed by the resistors Rf and
Re.
Since the open loop gain A is not infinite, the resistor
Rp and the input impedance Rin must be consid-
ered.Hence the voltage gain is
G = − Rf
Re x
1
1 + 1
A (1 +
Rf
Rp ⁄ ⁄ Re ⁄ ⁄ Rin)
(2)
II.1.2.1 - Input voltage Vin < Vref (black picture)
In this case the current flowing in R7 and T1 de-
creases whilst the collector voltage of T4 and the
output voltage both increase. In the extreme case,
I(T1) = I(R7) = 0 and VO= VDD-VGS(T5)
In order to charge the tube capacitor the voltage is
fed to the cathode output in two ways:
- through the PMOS (with a VGS difference) for the
low frequency part
- through the capacitor C for the high frequency
part (output signal leading edge)
To correctly transmit the rising edge, the value of
the capacitor C must be high compared to CL.
With the current values used (C = 1nF,CL = 10pF),
the attenuation is very small (0.99)
II.1.2.2 - Input voltage Vin > Vref (white picture)
In this case,the current in R7 and T1 increases with
an accompanying drop of T4’s collector voltage until
T1 and T4 are saturated. At this point:
VO ≅ VC(T4) ≅ VCC
During a high to low transition (i.e. black-white
picture), the beam current is absorbed in two ways:
- through the capacitor C and the compound
PMOS T6-T7 for the high frequency part (falling
edge)
- through the PMOS T8 and the resistor Rm for the
low frequency part.
II.2 - Beam Current Monitoring
II.2.1 - Stationary state
The beam current monitoring is performed by the
PMOS T8 and the resistor Rm. When measuring low
currents (leakage, quasi cutoff),the Rm value is
generally high. When measuring high currents
(drive, average or peak beam current),Rm is gen-
erally bypassed by a lower impedance.
It should be noted that the current supplied by the
three guns flows through this resistor.Hence,with
too large a value for the resistor Rm,the cathode
voltage of the tubes will become too high for the
required operating current values.This is a funda-
mental difference between the TEA5101A and dis-
crete video amps. In discrete video amps, the
current monitoring transistor is a high voltage PNP
bipolar which may saturate. In this case the beam
current can flow through the transistor base and it
is no longer monitored by the video processor. This
effect does not occur with the TEA 5101A.
II.2.2 - Transient phase : low current measure-
ments
The cut-off adjustment sequence is generally as
follows:
In a first step, the cathode is set to a high voltage
(180V) in order to blank the CRT and to measure
the leakage current. In a second step, the tube is
slighly switched on to measure a very low current
(quasi cut-off current). This operation is performed
by setting the cathode voltage to about 150V and
adjusting it until the proper current is obtained. The
maximum time available to do this operation is
generally about 52µs.
Figure 3 shows the simplified diagram of the
TEA5101A output, the voltages during the different
steps,and the stationary state the system must
reach for correct adjustment.
I.1 - Voltage Amplifier
II.1.1 - Bias conditions Vin = Vref
The bias point is fixed by the feedback resistor
Rf,the bias resistor Rp, and by the internal refer-
ence voltage when Vin = Vref.
If VO is the output voltage (pin 9) :
VO = (1 + Rf
Rp) x Vref (1)
In this state T1 and T2 are conducting. A current
flows in R7 and T4 soT5 is on. The T5 drain current
is fed to the amplifier input through the feedback
resistor. The current in R7 is:
I(R7) = VDD − VO − VGS(T5)
R7
≅ VDD − VO
R7
and the current in T5 and Rf is :
I(T5) = VO − Vref
Rf
≅ VO
Rf
Thus the total current absorbed by each channel of
the TEA5101A is :
VDD
R7 + VO x (1
Rf − 1
R7)
The cathode (pin 7) output voltage is:
VO + VGS(T8) = VO
The beam current is absorbed by T8 and Rm. The
voltage developed across Rm by this current is fed
to the videoprocessor in order to monitor the beam
current.
II.1.2 - Dynamic operation
The TEA5101A operates as a closed loop amplifier,
with its voltage gain fixed by the resistors Rf and
Re.
Since the open loop gain A is not infinite, the resistor
Rp and the input impedance Rin must be consid-
ered.Hence the voltage gain is
G = − Rf
Re x
1
1 + 1
A (1 +
Rf
Rp ⁄ ⁄ Re ⁄ ⁄ Rin)
(2)
II.1.2.1 - Input voltage Vin < Vref (black picture)
In this case the current flowing in R7 and T1 de-
creases whilst the collector voltage of T4 and the
output voltage both increase. In the extreme case,
I(T1) = I(R7) = 0 and VO= VDD-VGS(T5)
In order to charge the tube capacitor the voltage is
fed to the cathode output in two ways:
- through the PMOS (with a VGS difference) for the
low frequency part
- through the capacitor C for the high frequency
part (output signal leading edge)
To correctly transmit the rising edge, the value of
the capacitor C must be high compared to CL.
With the current values used (C = 1nF,CL = 10pF),
the attenuation is very small (0.99)
II.1.2.2 - Input voltage Vin > Vref (white picture)
In this case,the current in R7 and T1 increases with
an accompanying drop of T4’s collector voltage until
T1 and T4 are saturated. At this point:
VO ≅ VC(T4) ≅ VCC
During a high to low transition (i.e. black-white
picture), the beam current is absorbed in two ways:
- through the capacitor C and the compound
PMOS T6-T7 for the high frequency part (falling
edge)
- through the PMOS T8 and the resistor Rm for the
low frequency part.
II.2 - Beam Current Monitoring
II.2.1 - Stationary state
The beam current monitoring is performed by the
PMOS T8 and the resistor Rm. When measuring low
currents (leakage, quasi cutoff),the Rm value is
generally high. When measuring high currents
(drive, average or peak beam current),Rm is gen-
erally bypassed by a lower impedance.
It should be noted that the current supplied by the
three guns flows through this resistor.Hence,with
too large a value for the resistor Rm,the cathode
voltage of the tubes will become too high for the
required operating current values.This is a funda-
mental difference between the TEA5101A and dis-
crete video amps. In discrete video amps, the
current monitoring transistor is a high voltage PNP
bipolar which may saturate. In this case the beam
current can flow through the transistor base and it
is no longer monitored by the video processor. This
effect does not occur with the TEA 5101A.
II.2.2 - Transient phase : low current measure-
ments
The cut-off adjustment sequence is generally as
follows:
In a first step, the cathode is set to a high voltage
(180V) in order to blank the CRT and to measure
the leakage current. In a second step, the tube is
slighly switched on to measure a very low current
(quasi cut-off current). This operation is performed
by setting the cathode voltage to about 150V and
adjusting it until the proper current is obtained. The
maximum time available to do this operation is
generally about 52µs.
Figure 3 shows the simplified diagram of the
TEA5101A output, the voltages during the different
steps,and the stationary state the system must
reach for correct adjustment.

I.1 - Voltage Amplifier
II.1.1 - Bias conditions Vin = Vref
The bias point is fixed by the feedback resistor
Rf,the bias resistor Rp, and by the internal refer-
ence voltage when Vin = Vref.
If VO is the output voltage (pin 9) :
VO = (1 + Rf
Rp) x Vref (1)
In this state T1 and T2 are conducting. A current
flows in R7 and T4 soT5 is on. The T5 drain current
is fed to the amplifier input through the feedback
resistor. The current in R7 is:
I(R7) = VDD − VO − VGS(T5)
R7
≅ VDD − VO
R7
and the current in T5 and Rf is :
I(T5) = VO − Vref
Rf
≅ VO
Rf
Thus the total current absorbed by each channel of
the TEA5101A is :
VDD
R7 + VO x (1
Rf − 1
R7)
The cathode (pin 7) output voltage is:
VO + VGS(T8) = VO
The beam current is absorbed by T8 and Rm. The
voltage developed across Rm by this current is fed
to the videoprocessor in order to monitor the beam
current.
II.1.2 - Dynamic operation
The TEA5101A operates as a closed loop amplifier,
with its voltage gain fixed by the resistors Rf and
Re.
Since the open loop gain A is not infinite, the resistor
Rp and the input impedance Rin must be consid-
ered.Hence the voltage gain is
G = − Rf
Re x
1
1 + 1
A (1 +
Rf
Rp ⁄ ⁄ Re ⁄ ⁄ Rin)
(2)
II.1.2.1 - Input voltage Vin < Vref (black picture)
In this case the current flowing in R7 and T1 de-
creases whilst the collector voltage of T4 and the
output voltage both increase. In the extreme case,
I(T1) = I(R7) = 0 and VO= VDD-VGS(T5)
In order to charge the tube capacitor the voltage is
fed to the cathode output in two ways:
- through the PMOS (with a VGS difference) for the
low frequency part
- through the capacitor C for the high frequency
part (output signal leading edge)
To correctly transmit the rising edge, the value of
the capacitor C must be high compared to CL.
With the current values used (C = 1nF,CL = 10pF),
the attenuation is very small (0.99)
II.1.2.2 - Input voltage Vin > Vref (white picture)
In this case,the current in R7 and T1 increases with
an accompanying drop of T4’s collector voltage until
T1 and T4 are saturated. At this point:
VO ≅ VC(T4) ≅ VCC
During a high to low transition (i.e. black-white
picture), the beam current is absorbed in two ways:
- through the capacitor C and the compound
PMOS T6-T7 for the high frequency part (falling
edge)
- through the PMOS T8 and the resistor Rm for the
low frequency part.
II.2 - Beam Current Monitoring
II.2.1 - Stationary state
The beam current monitoring is performed by the
PMOS T8 and the resistor Rm. When measuring low
currents (leakage, quasi cutoff),the Rm value is
generally high. When measuring high currents
(drive, average or peak beam current),Rm is gen-
erally bypassed by a lower impedance.
It should be noted that the current supplied by the
three guns flows through this resistor.Hence,with
too large a value for the resistor Rm,the cathode
voltage of the tubes will become too high for the
required operating current values.This is a funda-
mental difference between the TEA5101A and dis-
crete video amps. In discrete video amps, the
current monitoring transistor is a high voltage PNP
bipolar which may saturate. In this case the beam
current can flow through the transistor base and it
is no longer monitored by the video processor. This
effect does not occur with the TEA 5101A.
II.2.2 - Transient phase : low current measure-
ments
The cut-off adjustment sequence is generally as
follows:
In a first step, the cathode is set to a high voltage
(180V) in order to blank the CRT and to measure
the leakage current. In a second step, the tube is
slighly switched on to measure a very low current
(quasi cut-off current). This operation is performed
by setting the cathode voltage to about 150V and
adjusting it until the proper current is obtained. The
maximum time available to do this operation is
generally about 52µs.
Figure 3 shows the simplified diagram of the
TEA5101A output, the voltages during the different
steps,and the stationary state the system must
reach for correct adjustment.

During the blanking phase, the tube is switched off,
the PMOS is switched off and its VGS voltage is
equal to the pinch-off voltage (about 1.5V). The
voltages at the different nodes are shown in figure
3 (V(9) = 180V, V(k) = 181.5V). The falling edge of
the cutoff pulse is instantaneously transmitted by
the capacitor C. When the stationary state is
reached, the cathode voltage will be 152.5V if the
voltage on pin 9 is 150V, as the VGS voltage of the
conducting PMOS is about 2.5V.
We can see that the voltage
on C must increase by
an amount of ∆Vc = 1V. This charge is furnished by
the tube capacitor which is discharged by an
amount of ∆VCL = 29V with a time constant equal
to R x CL (10 ns). By considering the energy
balance, we can calculate the maximum charge
∆Vmax that CL can furnished to C
∆Vmax = √CL
C x ∆VCL ≅ 3V
Since this voltage is greater than ∆VC, the capacitor
C can be charged and the stationary state is
reached without any contribution being required
from the tube current,i.e. the whole tube current
can flow through the PMOS and the adjustment can
be performed correctly.
Considering higher voltage and beam current
swings, the margin is greater because:
- the voltage swing across the tube capacitor is
greater
- the tube current is higher and the picture is not
disturbed even if part of the beam current is used
to charge the capacitor C.