Richtige Fernseher haben Röhren!

Richtige Fernseher haben Röhren!

In Brief: On this site you will find pictures and information about some of the electronic, electrical and electrotechnical technology relics that the Frank Sharp Private museum has accumulated over the years .

Premise: There are lots of vintage electrical and electronic items that have not survived well or even completely disappeared and forgotten.

Or are not being collected nowadays in proportion to their significance or prevalence in their heyday, this is bad and the main part of the death land. The heavy, ugly sarcophagus; models with few endearing qualities, devices that have some over-riding disadvantage to ownership such as heavy weight,toxicity or inflated value when dismantled, tend to be under-represented by all but the most comprehensive collections and museums. They get relegated to the bottom of the wants list, derided as 'more trouble than they are worth', or just forgotten entirely. As a result, I started to notice gaps in the current representation of the history of electronic and electrical technology to the interested member of the public.

Following this idea around a bit, convinced me that a collection of the peculiar alone could not hope to survive on its own merits, but a museum that gave equal display space to the popular and the unpopular, would bring things to the attention of the average person that he has previously passed by or been shielded from. It's a matter of culture. From this, the Obsolete Technology Tellye Web Museum concept developed and all my other things too. It's an open platform for all electrical Electronic TV technology to have its few, but NOT last, moments of fame in a working, hand-on environment. We'll never own Colossus or Faraday's first transformer, but I can show things that you can't see at the Science Museum, and let you play with things that the Smithsonian can't allow people to touch, because my remit is different.

There was a society once that was the polar opposite of our disposable, junk society. A whole nation was built on the idea of placing quality before quantity in all things. The goal was not “more and newer,” but “better and higher" .This attitude was reflected not only in the manufacturing of material goods, but also in the realms of art and architecture, as well as in the social fabric of everyday life. The goal was for each new cohort of children to stand on a higher level than the preceding cohort: they were to be healthier, stronger, more intelligent, and more vibrant in every way.

The society that prioritized human, social and material quality is a Winner. Truly, it is the high point of all Western civilization. Consequently, its defeat meant the defeat of civilization itself.

Today, the West is headed for the abyss. For the ultimate fate of our disposable society is for that society itself to be disposed of. And this will happen sooner, rather than later.

OLD, but ORIGINAL, Well made, Funny, Not remotely controlled............. and not Made in CHINA.

How to use the site:

- If you landed here via any Search Engine, you will get what you searched for and you can search more using the search this blog feature provided by Google. You can visit more posts scrolling the left blog archive of all posts of the month/year,
or you can click on the main photo-page to start from the main page. Doing so it starts from the most recent post to the older post simple clicking on the Older Post button on the bottom of each page after reading , post after post.

You can even visit all posts, time to time, when reaching the bottom end of each page and click on the Older Post button.

- If you arrived here at the main page via bookmark you can visit all the site scrolling the left blog archive of all posts of the month/year pointing were you want , or more simple You can even visit all blog posts, from newer to older, clicking at the end of each bottom page on the Older Post button.
So you can see all the blog/site content surfing all pages in it.

- The search this blog feature provided by Google is a real search engine. If you're pointing particular things it will search IT for you; or you can place a brand name in the search query at your choice and visit all results page by page. It's useful since the content of the site is very large.

Note that if you don't find what you searched for, try it after a period of time; the site is a never ending job !

Every CRT Television saved let revive knowledge, thoughts, moments of the past life which will never return again.........

Many contemporary "televisions" (more correctly named as displays) would not have this level of staying power, many would ware out or require major services within just five years or less and of course, there is that perennial bug bear of planned obsolescence where components are deliberately designed to fail and, or manufactured with limited edition specificities..... and without considering........picture......sound........quality........

..............The bitterness of poor quality is remembered long after the sweetness of todays funny gadgets low price has faded from memory........ . . . . . .....
Don't forget the past, the end of the world is upon us! Pretty soon it will all turn to dust!

Have big FUN ! !

©2010, 2011, 2012, 2013, 2014 Frank Sharp - You do not have permission to copy photos and words from this blog, and any content may be never used it for auctions or commercial purposes, however feel free to post anything you see here with a courtesy link back, btw a link to the original post here , is mandatory.
All sets and apparates appearing here are property of
Engineer Frank Sharp. NOTHING HERE IS FOR SALE !

Wednesday, February 5, 2014


The TELEVIDEON MOD. TEE112  is a 12 inches (31cm) portable little B/W television with 8 programs keyboard preselection with potentiometric tuning system.
The mechanical turret approach to television tuning has been used almost exclusively for the past 60 years. Even though replete with the inherent disadvantages of mechanical complexity, unreliability and cost, such apparatus has been technically capable of performing its intended function and as a result the consumer has had to bear the burdens associated with the device. However, with the " recent "  Broadcast demands for parity of tuning for UHF and VHF channels, the increasing number of UHF and cable TV stations have imposed new tuning performance requirements which severely tax the capability of the mechanical turret tuner. Consequently, attempts are now being made to provide all electronic tuning to meet the new requirements.

 One " " new " " tuning system currently being incorporated in some television receivers uses a varactor tuner which overcomes some of the disadvantages of mechanical turret tuner by accomplishing tuning electronically. As the name indicates, the heart of such a tuner is a varactor diode which is used as a capacitive tuning element in the RF and local oscillator sections. In this system, channel selection is made by applying a given reverse bias voltage to the varactor to change its electrical capacitance. The channel selection biasing can be performed by mechanically or electrically switching approximately 5 or many more preset potentiometers.
The invention relates to a tuning unit with bandswitch for high frequency receivers, especially radio and television receivers, having a potentiometer system for the control of capacity diodes, the said potentiometer system consisting of a plurality of parallel resistance paths along which wiper contacts can be driven by means of screw spindles disposed adjacent one another in a common insulating material housing in which a bandswitch formed of metal rods is associated with each tuning spindle.
In these tuning units, the working voltages of the capacity diodes in the tuning circuits are recorded once a precise tuning to the desired frequency has been performed. A potentiometer tuning system has great advantages over the formerly used channel selectors operating with mechanically adjustable capacitors (tuning condensers) or mechanically adjustable inductances (variometers), mainly because it is not required to have such great precision in its tuning mechanism.
Tuning units with bandswitches formed of variable resistances and combined with interlocking pushbuttons controlling the supply of recorded working voltages to capacity diodes are known. Channel selection is accomplished by depressing the knobs, and the tuning or fine tuning are performed by turning the knobs. The resistances serving as voltage dividers in these tuning units are combined into a component unit such that they are in the form of a ladderlike pattern on a common insulating plate forming the cover of the housing in which the tuning spindles and wiper contacts corresponding to the variable resistances are housed. The number of resistances corresponds to the number of channels or frequencies which are to be recorded. The wiper contact picks up a voltage which, when applied to the capacity diodes determines their capacitance and hence the frequency of the corresponding oscillating circuit. The adjustment of the wipers is performed by turning the tuning spindle coupled to the tuning knob. By the depression of a button the electrical connection between a contact rod and a tuning spindle is brought about and thus the selected voltage is applied to the capacity diodes. Since the push buttons release one another, it is possible simply by depressing another button to tune to a different receiving frequency or a different channel, as the case may be.
The problem with such arrangement is that it quite seriously limits the number of channels available to the consumer. Additionally, it suffers from the drawback that all potentiometers require adjusting for the desired channels. The VHF channels are usually factory adjusted while the six UHF channels require on-location adjustment. Moreover, using this arrangement, the only indication--during adjustment--of which channel is selected is by station identification.
All commands are above the cabinet located and was possible to applicate a plastic front screen over the crt face.
It has a Transistorized horizontal deflection circuits  made up of a horizontal switching or output transistor, a diode, one or more capacitors and a deflection winding. The output transistor, operating as a switch, is driven by a horizontal rate square wave signal and conducts during a portion of the horizontal trace interval. A diode, connected in parallel with the transistor, conducts during the remainder of the trace interval. A retrace capacitor and the deflection yoke winding are coupled in parallel across the transistor-diode combination. Energy is transferred into and out of the deflection winding via the diode and output transistor during the trace interval and via the retrace capacitor during the retrace interval.
In some television receivers, the collector of the horizontal output transistor is coupled to the B+ power supply through the primary windings of the high voltage transformer.

Televideon - Radio and Television; Italy   was a manufacturer of record player and tvs. The brand was from the FORMENTI group.

Television set kindly donated to me by Marshal Elia Z.

FORMENTI GROUP HISTORY It was founded in Desio ( Milan ) from Carlo Formenti in 1947 , under the name Carlo Formenti sas, and began his activities with the construction of radio , designed by the same founder, private label and under contract with other brands.

In 1953 the activities were moved to a new factory with 80 employees and was launched early production of TV sets in black and white , mostly as a subcontractor for major brands like Telefunken , Singer and Siemens .

Three years later began exports to foreign countries, and always as a subcontractor produced devices with trademarks for the large retail chains in Europe.

In 1961 took over the Italian plant of U.S. Admiral of Concorezzo with its 1,000 employees and a production capacity of 500 units per day, the license for the brand , and continued with the production of television and radio.
Admiral was formed Formenti SpA based in Milan , Lombardy and the company began its period of expansion so that within a few years led her to be one of the largest in the industry nationally.

Moreover, the production was extended to the field of "white" with the start of production of washing machines .

During the sixties , the head office and administration of the company, which changed its name to Industrie Formenti SpA Italy, was moved to Liss .

In 1972 was inaugurated a new plant in Sessa Aurunca for the production of color televisions with brands Admiral, Phoenix , Philco , and Televideon Uranya. Formenti was actually the first in Italy to produce the equipment, but until 1976 were marketed exclusively in the Central European countries, especially France and Germany , where for some time had already begun broadcasting in color.

Since 1983 the Group Formenti appeared on the market with another brand, White-Westinghouse , which became the most important with TVs, VCRs and household products from the same company Brianza.
In subsequent years Formenti used other brands as DuMont and Schaub-Lorenz .

Partially affected by the general crisis of the Italian electronic, in 1984 the public financial REL entered the capital Formenti record the 27.75% of the shares . The company, which by then had 580 employees scattered in the two plants, however, was one of the few entries in orbit REL able to repair their accounts in a short time.

In 1989 was started the modernization and extension of the establishment of Sessa, the installation of assembly automation , which allowed the company to undergo an increase in productivity from 300 to 2,200 pieces per day,The establishment of Concorezzo was instead intended to produce a new monitor and display for professional computer and ATM .

The company's continued A successful period during the nineties . Its sales went from over 108 billion lire in 1991 to 310 made in 1999 . Meanwhile, the Group acquired it in Monza 1997 the brand Imperial and the failed Sèleco , noting the brands Brionvega and Tandberg and the establishment of Pordenone (activated in 1998 ), and finally the Formenti family formed a new group called Sèleco-Formenti SpA, the which Charles, the founder, assumed the presidency, while the sons John and Julius, held the positions of CEO and director, respectively .

But since the year 2000 for Formenti begins a period of severe crisis. The company's balance sheet records a large deficit due to declining sales, the latter factor caused by the aggressive competition of producers Turkish color TV, put on the market at competitive prices clearly and without adequate anti- dumping on the part of ' European Union .

Also as a result of the difficulties facing the company, it decided to close the two plants Concorezzo ( 2000 ) and Sessa Aurunca ( 2003 ), respectively, 130 and 440 employees, as well as staff reductions in the Friuli .

In 2004 the company was put into liquidation , and then it is decided by the commissioner, the Court of Monza that appointment as curator Professor. Francesco Fimmanò. After a long period of unemployment benefits for workers of the company, shed of Pordenone and brands Sèleco, Brionvega and Imperial in 2006 are awarded to the Super / / Fluo , a company based in Udine owned by the brothers Carlo and Marco Asquini. The new property will only tempoi to produce some brand Imperial model tv, radio, and a Seleco brand new LCD TV brand Brionvega before the bankruptcy of the same. From this to save the brand Super / / Fluo, while the other brands owned and acquired by Formenti will pass to the new property called Selek Brionvega Technology and the mark for both audio and video production will switch to SIM2 Multimedia, the renowned Italian company field projectors.

Today  this industry and brand are DEAD !!!


The tuning circuits has a large knob potentiometers tuning system which use voltage controlled capacitances such as varactor diodes as the frequency determining elements.

Therefore a stable AFC circuit is developed:

A superheterodyne receiver having an automatic intermediate frequency control circuit with means to prevent the faulty regulation thereof. The receiver has means for receiving a radio frequency signal and mixing the same with the output of a superheterodyne oscillator. This produces an intermediate frequency signal which is coupled to a frequency or phase discriminator to produce an error signal for controlling the frequency of the superheterodyne oscillator. A regulation circuit is provided having an electronic switch to interrupt the feedback circuit when only unwanted frequencies tend to produce faulty regulation of the superheterodyne oscillator.

Power supply is realized with mains transformer and Linear transistorized power supply stabilizer, A DC power supply apparatus includes a rectifier circuit which rectifies an input commercial AC voltage. The rectifier output voltage is smoothed in a smoothing capacitor. Voltage stabilization is provided in the stabilizing circuits by the use of Zener diode circuits to provide biasing to control the collector-emitter paths of respective transistors.A linear regulator circuit according to an embodiment of the present invention has an input node receiving an unregulated voltage and an output node providing a regulated voltage. The linear regulator circuit includes a voltage regulator, a bias circuit, and a current control device.

In one embodiment, the current control device is implemented as an NPN bipolar junction transistor (BJT) having a collector electrode forming the input node of the linear regulator circuit, an emitter electrode coupled to the input of the voltage regulator, and a base electrode coupled to the second terminal of the bias circuit. A first capacitor may be coupled between the input and reference terminals of the voltage regulator and a second capacitor may be coupled between the output and reference terminals of the voltage regulator. The voltage regulator may be implemented as known to those skilled in the art, such as an LDO or non-LDO 3-terminal regulator or the like.
The bias circuit may include a bias device and a current source. The bias device has a first terminal coupled to the output terminal of the voltage regulator and a second terminal coupled to the control electrode of the current control device. The current source has an input coupled to the first current electrode of the current control device and an output coupled to the second terminal of the bias device. A capacitor may be coupled between the first and second terminals of the bias device.
In the bias device and current source embodiment, the bias device may be implemented as a Zener diode, one or more diodes coupled in series, at least one light emitting diode, or any other bias device which develops sufficient voltage while receiving current from the current source. The current source may be implemented with a PNP BJT having its collector electrode coupled to the second terminal of the bias device, at least one first resistor having a first end coupled to the emitter electrode of the PNP BJT and a second end, a Zener diode and a second resistor. The Zener diode has an anode coupled to the base electrode of the PNP BJT and a cathode coupled to the second end of the first resistor. The second resistor has a first end coupled to the anode of the Zener diode and a second end coupled to the reference terminal of the voltage regulator. A second Zener diode may be included having an anode coupled to the cathode of the first Zener diode and a cathode coupled to the first current electrode of the current control device.
A circuit is disclosed for improving operation of a linear regulator, having an input terminal, an output terminal, and a reference terminal. The circuit includes an input node, a transistor, a bias circuit, and first and second capacitors. The transistor has a first current electrode coupled to the input node, a second current electrode for coupling to the input terminal of the linear regulator, and a control electrode. The bias circuit has a first terminal for coupling to the output terminal of the linear regulator and a second terminal coupled to the control electrode of the transistor. The first capacitor is for coupling between the input and reference terminals of the linear regulator, and the second capacitor is for coupling between the output and reference terminals of the linear regulator. The bias circuit develops a voltage sufficient to drive the control terminal of the transistor and to operate the linear regulator. The bias circuit may be a battery, a bias device and a current source, a floating power supply, a charge pump, or any combination thereof. The transistor may be implemented as a BJT or FET or any other suitable current controlled device.

Power Supply: The examples chosen are taken from manufacturers' circuit diagrams and are usually simplified to emphasise the fundamental nature of the circuit. For each example the particular transistor properties that are exploited to achieve the desired performance are made clear. As a rough and ready classification the circuits are arranged in order of frequency: this part is devoted to circuits used at zero frequency, field frequency and audio frequencies. Series Regulator Circuit Portable television receivers are designed to operate from batteries (usually 12V car batteries) and from the a.c. mains. The receiver usually has an 11V supply line, and circuitry is required to ensure that the supply line is at this voltage whether the power source is a battery or the mains. The supply line also needs to have good regulation, i.e. a low output resistance, to ensure that the voltage remains constant in spite of variations in the mean current taken by some of the stages in the receiver. Fig. 1 shows a typical circuit of the power -supply arrangements. The mains transformer and bridge rectifier are designed to deliver about 16V. The battery can be assumed to give just over 12V. Both feed the regulator circuit Trl, Tr2, Tr3, which gives an 11V output and can be regarded as a three -stage direct -coupled amplifier. The first stage Tr 1 is required to give an output current proportional to the difference between two voltages, one being a constant voltage derived from the voltage reference diode D I (which is biased via R3 from the stabilised supply). The second voltage is obtained from a preset potential divider connected across the output of the unit, and is therefore a sample of the output voltage. In effect therefore Tr 1 compares the output voltage of the unit with a fixed voltage and gives an output current proportional to the difference between them. Clearly a field-effect transistor could do this, but the low input resistance of a bipolar transistor is no disadvantage and it can give a current output many times that of a field-effect transistor and is generally preferred therefore. The output current of the first stage is amplified by the two subsequent stages and then becomes the output current of the unit. Clearly therefore Tr2 and Tr3 should be current amplifiers and they normally take the form of emitter followers or common emitter stages (which have the same current gain). By adjusting the preset control we can alter the fraction of the output voltage' applied to the first stage and can thus set the output voltage of the unit at any desired value within a certain range. By making assumptions about the current gain of the transistors we can calculate the degree of regulation obtainable. For example, suppose the gain of Tr2 and Tr3 in cascade is 1,000, and that the current output demanded from the unit changes by 0.1A (for example due to the disconnection of part of the load). The corresponding change in Tr l's collector current is 0.1mA and, if the standing collector current of Tr 1 is 1mA, then its mutual conductance is approximately 4OmA/V and the base voltage must change by 2.5mV to bring about the required change in collector current. If the preset potential divider feeds one half of the output voltage to Tr l's base, then the change in output voltage must be 5mV. Thus an 0.1A change in output current brings about only 5mV change in output voltage: this represents an output resistance of only 0.0552.

TDA1170 vertical deflection FRAME DEFLECTION INTEGRATED CIRCUITGENERAL DESCRIPTION f The TDA1170 and TDA1270 are monolithic integrated
circuits designed for use in TV vertical deflection systems. They are manufactured using
the Fairchild Planar* process.
Both devices are supplied in the 12-pin plastic power package with the heat sink fins bent
for inse
rtion into the printed circuit board.
The TDA1170 is designed primarily for large and small screen black and white TV
receivers and industrial TV monitors. The TDA1270 is designed primarily for driving
complementary vertical deflection output stages in color TV receivers and industrial
The vertical oscillator is directly synchronized by the sync pulses (positive or negative); therefore its free
running frequency must be lower than the sync frequency. The use of current feedback causes the yoke
current to be independent of yoke resistance variations due to thermal effects, Therefore no thermistor is
required in series with the yoke. The flyback generator applies a voltage, about twice the supply voltage, to
the yoke. This produces a short flyback time together with a high useful power to dissipated power


The basic essentials of a transistor line output stage are shown in Fig. 1(a). They comprise: a line output transformer which provides the d.c. feed to the line output transistor and serves mainly to generate the high -voltage pulse from which the e.h.t. is derived, and also in practice other supplies for various sections of the receiver; the line output transistor and its parallel efficiency diode which form a bidirectional switch; a tuning capacitor which resonates with the line output transformer primary winding and the scan coils to determine the flyback time; and the scan coils, with a series capacitor which provides a d.c. block and also serves to provide slight integration of the deflection current to compensate for the scan distortion that would otherwise be present due to the use of flat screen, wide deflection angle c.r.t.s. This basic circuit is widely used in small -screen portable receivers with little elaboration - some use a pnp output transistor however, with its collector connected to chassis.

Circuit Variations:
Variations to the basic circuit commonly found include: transposition of the scan coils and the correction capacitor; connection of the line output transformer primary winding and its e.h.t. overwinding in series; connection of the deflection components to a tap on the transformer to obtain correct matching of the components and conditions in the stage; use of a boost diode which operates in identical manner to the arrangement used in valve line output stages, thereby increasing the effective supply to the stage; omission of the efficiency diode where the stage is operated from an h.t. line, the collector -base junction of the line output transistor then providing the efficiency diode action without, in doing so, producing scan distortion; addition of inductors to provide linearity and width adjustment; use of a pair of series -connected line output transistors in some large -screen colour chassis; and in colour sets the addition of line convergence circuitry which is normally connected in series between the line scan coils and chassis. These variations on the basic circuit do not alter the basic mode of operation however.

The most important fact to appreciate about the circuit is that when the transistor and diode are cut off during the flyback period - when the beam is being rapidly returned from the right-hand side of the screen to the left-hand side the tuning capacitor together with the scan coils and the primary winding of the line output transformer form a parallel resonant circuit: the equivalent circuit is shown in Fig. 1(b). The line output transformer primary winding and the tuning capacitor as drawn in Fig. 1(a) may look like a series tuned circuit, but from the signal point of view the end of the transformer primary winding connected to the power supply is earthy, giving the equivalent arrangement shown in Fig. 1(b).

The Flyback Period:
Since the operation of the circuit depends mainly upon what happens during the line flyback period, the simplest point at which to break into the scanning cycle is at the end of the forward scan, i.e. with the beam deflected to the right-hand side of the screen, see Fig. 2. At this point the line output transistor is suddenly switched off by the squarewave drive applied to its base. Prior to this action a linearly increasing current has been flowing in the line output transformer primary winding and the scan coils, and as a result magnetic fields have been built up around these components. When the transistor is switched off these fields collapse, maintaining a flow of current which rapidly decays to zero and returns the beam to the centre of the screen. This flow of current charges the tuning capacitor, and the voltage at A rises to a high positive value - of the order of 1- 2k V in large -screen sets, 200V in the case of mains/battery portable sets. The energy in the circuit is now stored in the tuning capacitor which next discharges, reversing the flow of current in the circuit with the result that the beam is rapidly deflected to the left-hand side of the screen - see Fig. 3. When the tuning capacitor has discharged, the voltage at A has fallen to zero and the circuit energy is once more stored in the form of magnetic fields around the inductive components. One half -cycle of oscillation has occurred, and the flyback is complete.

Energy Recovery:
First Part of Forward Scan The circuit then tries to continue the cycle of oscillation, i.e. the magnetic fields again collapse, maintaining a current flow which this time would charge the tuning capacitor negatively (upper plate). When the voltage at A reaches about -0.6V however the efficiency diode becomes forward biased and switches on. This damps the circuit, preventing further oscillation, but the magnetic fields continue to collapse and in doing so produce a linearly decaying current flow which provides the first part of the forward scan, the beam returning towards the centre of the screen - see Fig. 4. The diode shorts out the tuning capacitor but the scan correction capacitor charges during this period, its right-hand plate becoming positive with respect to its left-hand plate, i.e. point A. Completion of Forward Scan When the current falls to zero, the diode will switch off. Shortly before this state of affairs is reached however the transistor is switched on. In practice this is usually about a third of the way through the scan. The squarewave applied to its base drives it rapidly to saturation, clamping the voltage at point A at a small positive value - the collector emitter saturation voltage of the transistor. Current now flows via the transistor and the primary winding of the line output transformer, the scan correction capacitor discharges, and the resultant flow of current in the line scan coils drives the beam to the right-hand side of the screen see Fig. 5.

The transistor is then cut off again, to give the flyback, and the cycle of events recurs. The efficiency of the circuit is high since there is negligible resistance present. Energy is fed into the circuit in the form of the magnetic fields that build up when the output transistor is switched on. This action connects the line output transformer primary winding across the supply, and as a result a linearly increasing current flows through it. Since the width is
dependent on the supply voltage, this must be stabilised.

Harmonic Tuning:
There is another oscillatory action in the circuit during the flyback period. The considerable leakage inductance between the primary and the e.h.t. windings of the line output transformer, and the appreciable self -capacitance present, form a tuned circuit which is shocked into oscillation by the flyback pulse. Unless this oscillation is controlled, it will continue into and modulate the scan. The technique used to overcome this effect is to tune the leakage inductance and the associated capacitance to an odd harmonic of the line flyback oscillation frequency. By doing this the oscillatory actions present at the beginning of the scan cancel. Either third or fifth harmonic tuning is used. Third harmonic tuning also has the effect of increasing the amplitude of the e.h.t. pulse, and is generally used where a half -wave e.h.t. rectifier is employed. Fifth harmonic tuning results in a flat-topped e.h.t. pulse, giving improved e.h.t. regulation, and is generally used where an e.h.t. tripler is employed to produce the e.h.t. The tuning is mainly built into the line output transformer, though an external variable inductance is commonly found in colour chassis so that the tuning can be adjusted. With a following post I will go into the subject of modern TV line timebases in greater detail with other models and technology shown here at  Obsolete Technology Tellye !

RCA CA920 CA920AE / TBA920 line oscillator combination
The line oscillator combination CA920 TBA920 is a monolithic
integrated circuit intended for the horizontal deflection of the black and white
and colour TV sets
picture tube.


In a conventional line scan circuit, such as in the Plessey or Mullard TBA920 integrated circuit, a phase-locked oscillator is synchronized with the incoming horizontal synchronization (H-sync) pulses to provide a control signal to the line output circuit. This ensures that oscillation is maintained in the absence of the H-sync signal, for example, during maintenance in the case of a display terminal, or, in the case of a TV receiver, when untuned, and noise on the incoming signal is smoothed out. The phase relationship between the output and input is very precisely maintained to prevent line tearing and other picture defects.

Flyback voltage pulses occur at the terminals of the yoke of the cathode ray tube (CRT) between successive horizontal scans of the electron beam while the screen is blanked. Due to component tolerances and temperature effects, the flyback voltage pulses may be broadened and unless the broadened pulse falls within the blanking period, picture defects can occur. Conventionally, this problem has been overcome by designing the CRT circuits to allow a broad blanking period between the line scans: however, this has a deleterious effect on the efficiency of the display.
The design of the driver and horizontal output transistor stages is very critical. In typical circuits the horizontal output transistor conducts during the last half of the trace interval and is sharply cut-off to initiate retrace. During retrace a large retrace or flyback voltage appears at the collector of the horizontal output transistor. If the transistor is conducting when this voltage appears, the retrace current will be altered and the transistor will dissipate excessive power which generally will result in damage to the transistor. Since the horizontal output transistor is conducting heavily immediately before retrace is initiated, the stored charge in the base and collector regions of the transistor must be removed before the transistor will cease conducting collector current. Normally this stored charge is removed by applying a negative pulse to the base of an NPN transistor to cause a reverse base current to flow. This negative pulse, however, should be applied only as long as necessary to alleviate the voltage stress on the collector-base junction during retrace. R. J. Walker and R. Yu, "Horizontal Output Transistor Base Circuit Design," IEEE Transactions on Broadcast and Television Receivers, Vol. BTR-20, Aug. 1974, pp. 185-192, describe the various design requirements for the base circuit of the horizontal output transistor.
The above design requirements place severe limitations on the horizontal driver transistor and transformer design. In typical circuits conduction by the driver transistor during retrace and the first half of trace switches the output transistor off. The driver transformer stores energy during conduction by the driver transistor which is used to provide forward base current to the horizontal output transistor during the following conduction interval of the output transistor. Accordingly, the drive current in the primary winding must be of the proper amplitude and duration to maintain the horizontal output transistor in conduction in saturation during the second half of trace. When the driver transistor turns on again, the initial current flow through the primary winding of the driver transformer provides a turn-off pulse to the base of the output transistor. The amplitude and duration of the turn-off pulse are dependent on the inductances and the turns ratio of the driver transformer windings.

 THE TBA920 SYNC/TIMEBASE IC It has been quite common for some time for sync separation to be carried out in an i.c. but until 1971 this was as far as i.c.s had gone in television receiver timebase circuitry. With the recent introduction of the delta featured 110°  colour series however i.c.s have gone a step farther since this chassis uses a TBA920 as sync separator and line generator. A block diagram of this PHILIPS /Mullard  i.c. is shown in Fig. 1.
The video signal at about 2-7V peak -peak is fed to the sync separator section at pin 8, the composite sync waveform appearing at pin 7.
The noise gate switches off the sync separator when a positive -going input pulse is fed in at pin 9, an external noise limiter circuit being required .
The line sync pulses are shaped by R1 /C1 /C2/R2 and fed in to the oscillator phase detector section at pin 6.
The line oscillator waveform is fed internally to the oscillator phase detector circuit which produces at pin 12 a d.c. potential which is used to lock the line oscillator to the sync pulse frequency, the control potential being fed in at pin 15. The oscillator itself is a CR type whose waveform is produced by the charge and discharge of the external capacitor (C7) connected to pin 14. The oscillator frequency is set basically by C7 and R6 and can be varied by the control potential appearing at pin 15 from pin 12 and the external line hold control. Internally the line oscillator feeds a triangular waveform to the oscillator and flyback phase detector sections and the pulse width control section. The coincidence detector section is used to set the time constant of the oscillator phase detector circuit. It is fed internally with sync pulses from the sync separator section, and with line flyback pulses via pin 5. When the flyback pulses are out of phase with the sync pulses the impedance looking into pin 11 is high (21(Q). When the pulses are coincident the impedance falls to about 150Q and the oscillator phase detector circuit is then slow acting. The effect of this is to give fast pull -in when the pulses are out of sync and good noise immunity when they are in sync. The coincidence detector is controlled by the voltage on pin 10. When the sync and flyback pulses are in sync C3 is charged: when they are out of sync C3 discharges via R3. VTR use has been taken into consideration here. With a video recorder it is necessary to be able to follow the sync pulse phase variations that occur as a result of wow and flutter in the tape transport system, while noise is much less of a problem. For use with a VTR therefore the network on pin 10 can simply be left out so that the oscillator phase detector circuit is always fast acting. A second control loop is used to adjust the timing of the pulse output obtained from pin 2 to take into account the delay in the line output stage. The fly back phase detector compares the frequency of the flyback pulses fed in at pin 5 with the oscillator signal which has already been synchronised to the sync pulse frequency.
Any phase difference results in an output from pin 4 which is integrated and fed into the pulse width control section at pin 3. The potential at pin 3 sets the width of the output pulse obtained at pin 2: with a high positive voltage (via R11 and R12) at pin 3 a 1:1 mark -space ratio out- put pulse (32/us on, 32/us off) will be produced while a low potential at pin 3 (negative output at pin 4) will give a 16us output pulse at  the same frequency. The action of this control loop continues until the fly- back pulses are in phase with a fixed point on the oscillator waveform: the flyback pulses are then in phase with the sync pulses and delays in the line output stage are compensated. The output obtained at pin 2 is of low impedance and is suitable for driving valves, transistors or thyristors: R9 is necessary to provide current limiting.


Tuesday, February 4, 2014


The PHONOLA (PHILIPS) 37KV1232  is a 15 inches color television with 50 programs with vst tuning search and osd.On screen display (OSD) arrangements employed in video processing systems include a switching (or "multiplexing") network for switching between graphic image representative signals and normal video signals so that a graphic image can be displayed on the screen of a picture reproduction device either in place of the image represented by the video signals or together with (inserted in) the image. The graphic image can take the form of alphanumeric symbols or-pictorial graphics, and can be used to indicate status information, such as channel numbers or time, or operating instructions.

In an OSD arrangement for use in an analog video signal processing system, the multiplexing network typically operates to switch in levels corresponding to the desired intensity of respective portions of the graphic image at the time the graphic image portions are to be displayed. In such an arrangement the graphic image representative signals take the form of timing pulses which occur when the graphic image portions are to be displayed and are used to control the multiplexing network. Such an analog OSD arrangement can also be used in a digital video processing system, but requires that the video signals be first converted to analog form. While digital video signal processing systems typically include a digital-to-analog converter section in which the digital video signals are converted to analog form, it may be more cost effective for the OSD arrangement to be incorporated as an integral part of the digital video processing section.

The set is a simple color tv and was last Phonola / Philips model without AV Scart socket.

This models series was featuring the GR1AX a small monocarrier tv chassis briefly used before other types with SMPS isolated from mains since the GR1AX has direct mains SMPS supply.

Pictures are very bright and powerful.

Koninklijke Philips Electronics N.V. (Royal Philips Electronics Inc.), most commonly known as Philips, (Euronext: PHIA, NYSE: PHG) is a multinational Dutch electronics corporation.

Philips is one of the largest electronics companies in the world. In 2009, its sales were €23.18 billion. The company employs 115,924 people in more than 60 countries.

Philips is organized in a number of sectors: Philips Consumer Lifestyles (formerly Philips Consumer Electronics and Philips Domestic Appliances and Personal Care), Philips Lighting and Philips Healthcare (formerly Philips Medical Systems).
The company was founded in 1891 by Gerard Philips, a maternal cousin of Karl Marx, in Eindhoven, Netherlands. Its first products were light bulbs and other electro-technical equipment. Its first factory survives as a museum devoted to light sculpture. In the 1920s, the company started to manufacture other products, such as vacuum tubes (also known worldwide as 'valves'), In 1927 they acquired the British electronic valve manufacturers Mullard and in 1932 the German tube manufacturer Valvo, both of which became subsidiaries. In 1939 they introduced their electric razor, the Philishave (marketed in the USA using the Norelco brand name).
Philips was also instrumental in the revival of the Stirling engine.

As a chip maker, Philips Semiconductors was among the Worldwide Top 20 Semiconductor Sales Leaders.

In December 2005 Philips announced its intention to make the Semiconductor Division into a separate legal entity. This process of "disentanglement" was completed on 1 October 2006.

On 2 August 2006, Philips completed an agreement to sell a controlling 80.1% stake in Philips Semiconductors to a consortium of private equity investors consisting of Kohlberg Kravis Roberts & Co. (KKR), Silver Lake Partners and AlpInvest Partners. The sale completed a process, which began December 2005, with its decision to create a separate legal entity for Semiconductors and to pursue all strategic options. Six weeks before, ahead of its online dialogue, through a letter to 8,000 of Philips managers, it was announced that they were speeding up the transformation of Semiconductors into a stand-alone entity with majority ownership by a third party. It was stated then that "this is much more than just a transaction: it is probably the most significant milestone on a long journey of change for Philips and the beginning of a new chapter for everyone – especially those involved with Semiconductors".

In its more than 115 year history, this counts as a big step that is definitely changing the profile of the company. Philips was one of few companies that successfully made the transition from the electrical world of the 19th century into the electronic age, starting its semiconductor activity in 1953 and building it into a global top 10 player in its industry. As such, Semiconductors was at the heart of many innovations in Philips over the past 50 years.

Agreeing to start a process that would ultimately lead to the decision to sell the Semiconductor Division therefore was one of the toughest decisions that the Board of Management ever had to make.

On 21 August 2006, Bain Capital and Apax Partners announced that they had signed definitive commitments to join the expanded consortium headed by KKR that is to acquire the controlling stake in the Semiconductors Division.

On 1 September 2006, it was announced in Berlin that the name of the new semiconductor company founded by Philips is NXP Semiconductors.

Coinciding with the sale of the Semiconductor Division, Philips also announced that they would drop the word 'Electronics' from the company name, thus becoming simply Koninklijke Philips N.V. (Royal Philips N.V.).


The foundations of Philips were laid in 1891 when Anton and Gerard Philips established Philips & Co. in Eindhoven, the Netherlands. The company begun manufacturing carbon-filament lamps and by the turn of the century, had become one of the largest producers in Europe. Stimulated by the industrial revolution in Europe, Philips’ first research laboratory started introducing its first innovations in the x-ray and radio technology. Over the years, the list of inventions has only been growing to include many breakthroughs that have continued to enrich people’s everyday lives.

In the early years of Philips &; Co., the representation of the company name took many forms: one was an emblem formed by the initial letters of Philips ; Co., and another was the word Philips printed on the glass of metal filament lamps.

One of the very first campaigns was launched in 1898 when Anton Philips used a range of postcards showing the Dutch national costumes as marketing tools. Each letter of the word Philips was printed in a row of light bulbs as at the top of every card. In the late 1920s, the Philips name began to take on the form that we recognize today.

The now familiar Philips waves and stars first appeared in 1926 on the packaging of miniwatt radio valves, as well as on the Philigraph, an early sound recording device. The waves symbolized radio waves, while the stars represented the ether of the evening sky through which the radio waves would travel.

In 1930 it was the first time that the four stars flanking the three waves were placed together in a circle. After that, the stars and waves started appearing on radios and gramophones, featuring this circle as part of their design. Gradually the use of the circle emblem was then extended to advertising materials and other products.

At this time Philips’ business activities were expanding rapidly and the company wanted to find a trademark that would uniquely represent Philips, but one that would also avoid legal problems with the owners of other well-known circular emblems. This wish resulted in the combination of the Philips circle and the wordmark within the shield emblem.

In 1938, the Philips shield made its first appearance. Although modified over the years, the basic design has remained constant ever since and, together with the wordmark, gives Philips the distinctive identity that is still embraced today.

The first steps of CRT production by Philips started in the thirties with the Deutsche Philips Electro-Spezial gesellschaft in Germany and the Philips NatLab (Physics laboratory) in Holland. After the introduction of television in Europe, just after WWII there was a growing demand of television sets and oscilloscope equipment. Philips in Holland was ambitious and started experimental television in 1948. Philips wanted to be the biggest on this market. From 1948 there was a small Philips production of television and oscilloscope tubes in the town of Eindhoven which soon developed in mass production. In 1976 a part of the Philips CRT production went to the town of Heerlen and produced its 500.000'th tube in 1986. In 1994 the company in Heerlen changed from Philips into CRT-Heerlen B.V. specialized in the production of small monochrome CRT's for the professional market and reached 1.000.000 produced tubes in 1996. In this stage the company was able to produce very complicated tubes like storage CRT's.
In 2001 the company merged into Professional Display Systems, PDS worked on LCD and Plasma technology but went bankrupt in 2009. The employees managed a start through as Cathode Ray Technology which now in 2012 has to close it's doors due to the lack of sales in a stressed market. Their main production was small CRT's for oscilloscope, radar and large medical use (X-ray displays). New experimental developments were small Electron Microscopy, 3D-TV displays, X-Ray purposes and Cathode Ray Lithography for wafer production. Unfortunately the time gap to develop these new products was too big.

28 of September 2012, Cathode Ray Technology (the Netherlands), the last Cathode Ray Tube factory in Europe closed. Ironically the company never experienced so much publicity as now, all of the media brought the news in Holland about the closure. In fact this means the end of mass production 115 years after Ferdinand Braun his invention. The rapid introduction and acceptation of LCD and Plasma displays was responsible for a drastic decrease in sales. Despite the replacement market for the next couple of years in the industrial, medical and avionics sector.
The numbers are small and the last few CRT producers worldwide are in heavy competition.

Gerard Philips:

Gerard Leonard Frederik Philips (October 9, 1858, in Zaltbommel – January 27, 1942, in The Hague, Netherlands) was a Dutch industrialist, co-founder (with his father Frederik Philips) of the Philips Company as a family business in 1891. Gerard and his younger brother Anton Philips changed the business to a corporation by founding in 1912 the NV Philips' Gloeilampenfabrieken. As the first CEO of the Philips corporation, Gerard laid with Anton the base for the later Philips multinational.

Early life and education

Gerard was the first son of Benjamin Frederik David Philips (1 December 1830 – 12 June 1900) and Maria Heyligers (1836 – 1921). His father was active in the tobacco business and a banker at Zaltbommel in the Netherlands; he was a first cousin of Karl Marx.


Gerard Philips became interested in electronics and engineering. Frederik was the financier for Gerard's purchase of the old factory building in Eindhoven where he established the first factory in 1891. They operated the Philips Company as a family business for more than a decade.

Marriage and family

On March 19, 1896 Philips married Johanna van der Willigen (30 September 1862 – 1942). They had no children.

Gerard was an uncle of Frits Philips, whom he and his brother brought into the business. Later they brought in his brother's grandson, Franz Otten.

Gerard and his brother Anton supported education and social programs in Eindhoven, including the Philips Sport Vereniging (Philips Sports Association), which they founded. From it the professional football (soccer) department developed into the independent Philips Sport Vereniging N.V.

Anton Philips:

Anton Frederik Philips (March 14, 1874, Zaltbommel, Gelderland – October 7, 1951, Eindhoven) co-founded Royal Philips Electronics N.V. in 1912 with his older brother Gerard Philips in Eindhoven, the Netherlands. He served as CEO of the company from 1922 to 1939.

Early life and education

Anton was born to Maria Heyligers (1836 – 1921) and Benjamin Frederik David Philips (December 1, 1830 – June 12, 1900). His father was active in the tobacco business and a banker at Zaltbommel in the Netherlands. (He was a first cousin to Karl Marx.) Anton's brother Gerard was 16 years older.


In May 1891 the father Frederik was the financier and, with his son Gerard Philips, co-founder of the Philips Company as a family business. In 1912 Anton joined the firm, which they named Royal Philips Electronics N.V.

During World War I, Anton Philips managed to increase sales by taking advantage of a boycott of German goods in several countries. He provided the markets with alternative products.

Anton (and his brother Gerard) are remembered as being civic-minded. In Eindhoven they supported education and social programs and facilities, such as the soccer department of the Philips Sports Association as the best-known example.

Anton Philips brought his son Frits Philips and grandson Franz Otten into the company in their times. Anton took the young Franz Otten with him and other family members to escape the Netherlands just before the Nazi Occupation during World War II; they went to the United States. They returned after the war.

His son Frits Philips chose to stay and manage the company during the occupation; he survived several months at the concentration camp of Vught after his workers went on strike. He saved the lives of 382 Jews by claiming them as indispensable to his factory, and thus helped them evade Nazi roundups and deportation to concentration camps.

Philips died in Eindhoven in 1951.

Marriage and family

Philips married Anne Henriëtte Elisabeth Maria de Jongh (Amersfoort, May 30, 1878 – Eindhoven, March 7, 1970). They had the following children:

* Anna Elisabeth Cornelia Philips (June 19, 1899 – ?), married in 1925 to Pieter Franciscus Sylvester Otten (1895 – 1969), and had:
o Diek Otten
o Franz Otten (b. c. 1928 - d. 1967), manager in the Dutch electronics company Philips
* Frederik Jacques Philips (1905-2005)
* Henriëtte Anna Philips (Eindhoven, October 26, 1906 – ?), married firstly to A. Knappert (d. 1932), without issue; married secondly to G. Jonkheer Sandberg (d. September 5, 1935), without issue; and married thirdly in New York City, New York, on September 29, 1938 to Jonkheer Gerrit van Riemsdijk (Aerdenhout, January 10, 1911 – Eindhoven, November 8, 2005). They had the following children:
o ..., Jonkheerin Gerrit van Riemsdijk (b. Waalre, October 2, 1939), married at Waalre on February 17, 1968 to Johannes Jasper Tuijt (b. Atjeh, Koeta Radja, March 10, 1930), son of Jacobus Tuijt and wife Hedwig Jager, without issue
o ..., Jonkheerin Gerrit van Riemsdijk (b. Waalre, April 3, 1946), married firstly at Calvados, Falaise, on June 6, 1974 to Martinus Jan Petrus Vermooten (Utrecht, September 16, 1939 – Falaise, August 29, 1978), son of Martinus Vermooten and wife Anna Pieternella Hendrika Kwantes, without issue; married secondly in Paris on December 12, 1981 to Jean Yves Louis Bedos (Calvados, Rémy, January 9, 1947 – Calvados, Lisieux, October 5, 1982), son of Georges Charles Bedos and wife Henriette Louise Piel, without issue; and married thirdly at Manche, Sartilly, on September 21, 1985 to Arnaud Evain (b. Ardennes, Sedan, July 7, 1952), son of Jean Claude Evain and wife Flore Halleux, without issue
o ..., Jonkheerin Gerrit van Riemsdijk (b. Waalre, September 4, 1948), married at Waalre, October 28, 1972 to Elie Johan François van Dissel (b. Eindhoven, October 9, 1948), son of Willem Pieter
Jacob van Dissel and wife Francisca Frederike Marie Wirtz, without issue.

(To see the Internal Chassis Just click on Older Post Button on bottom page, that's simple !)




The TDA8305A is TV sub-system circuit, for colour television receivers with the following features.

 Vision IF amplifier with synchronous demodulator

Automatic gain control (AGC) detector suitable for negative modulation
AGC tuner

Automatic frequency control (AFC) circuit with sample-and-hold

Video preamplifier

Sound lF amplifier and demodulator

DC volume control or separate supply for starting the horizontal oscillator
Audio preamplifier

Horizontal synchronization circuit with two control loops

Vertical synchronization (divider system) and sawtooth generation with automatic amplitude
adjustment for 50 and 60 Hz

Transmitter identification (mute)

Generation of sandcastle pulse


Vision IF amplifier, demodulator and video amplifier

The IF amplifier of the TDA8305A has three AC~coupled stages, each stage having a control range
that exceeds 20 dB. AC-coupling means that the DC-feedback circuitry of the ampiifier (present in
the TDA4505) can be omitted, resulting in a saving of one pin. An additional advantage is the
symmetry of the amplifier which results in a less critical application.

In the TDA8305A the regenerated carrier signal islimited by a logarithmic limiter circuit before it is
passed on to a passive synchronous demodulator. The limiter has a very low differential phase shift
which results in good differential gain and phase figures.

The TDA8305A's video amplifier has a higher bandwidth and better linearity compared with that of
the TDA4505. A noise clamp is included in the video amplifier that limits the interference pulses
to a level just below the top sync. This circuit is more effective than the noise inverter used in the
TDA4505 and results in an improved picture stability, with respect to interference.

AFC circuit

In the TDA4505 and TDA8305A, the reference signal for the AFC circuit is obtained from the
demodulator tuned circuit which means only one tuned circuit and adjustment are needed. The
disadvantage with this method is that the frequency spectrum fed to the detector is determined by the
SAW filter characteristic. This spectrum is asymmetrical with respect to the picture carrier so that the
AFC output voltage is dependent on the video signal. This was the main problem found with the
TDA4505's AFC circuit.

To remove this problem the TDA8305A is equipped with a sample-and-hold circuit which samples
during the sync level of the signal. This means that only the carrier signal is available to the AFC
and it will not be affected by the video information. The additional pin required for this circuit is
provided by the pin that became available when the DC feedback circuit was removed from the IF
amplifier (see previous section).

Weak input signals will cause the drive signal of the AFC to contain a lot of noise. This noise signal
has an asymmetrical frequency spectrum that causes an offset in the AFC output voltage, this
offset can be reduced by applying a notch to the demodulator circuit. The sample-and-hold circuit
is followed by a high output impedance amplifier, therefore the AFC's control steepness is
dependent on the load impedance.

AGC circuit

The TDA8305A's AGC detector differs from that of the TDA4505 in that it doesn’t need the charge
resistor but has an internal current source. Also the circuitry between the detector capacitor and the
control stages has been changed to improve the signal-to-noise ratio of the video output signal

(no dips in the S/N ratio depending on the input signal amplitude). The point of tuner take-over
is preset by the voltage level at pin 1.

Sound circuit and horizontal oscillator starting function:

The input to the sound IF amplifier is by means of a bandpass filter coupling from the video output
(pin 17). The sound is demodulated and passed via a volume control stage to the audio

output amplifier. Volume control is obtained by connecting a potentiometer (10 k9.) between pin 11
and ground, or by supplying pin 11 with a variable voitage. Sound output is suppressed by an
internal mute signal when no TV signal is identified.

Improvement of sound quality was one of the main reasons for redesigning the TDA4505. To obtain
a better idea of the performance of the various circuits of the TDA4505 the following measurements
were carried out:
 Weak signal performance when a TBA120 is driven with an intercarrier signal obtained from the
vision IF circuit of the TDA4505 (the sound IF of the latter was not used)

 The same measurement for the sound IF circuit of the TDA4505 driven from another TDA4505
(again without using the sound IF circuit)

 The same measurement as in the first case but with the sound IF of the TDA4505 connected

From the results of these measurements it was established that the sound problem was caused by an
interaction between vision IF and sound circuits. The improved sound quality of the TDA8305A as
compared to the TDA4505 was achieved by:

 A very symmetrical vision-IF amplifier which is less sensitive to radiation from the sound IF

0 A change to the internal ground and supply connections of the IC to reduce coupling between
both circuits

DC volume control/Horizontal oscillator start

Horizontal oscillator; the operation depending on the application. During switchon if no current is
supplied to pin 11 this pin will act as a volume control. When a current of 9.0 mA is supplied to
pin 11 the volume control is set to a fixed output signal and the device will generate drive pulses
for the horizontal deflection. The main supply can then be derived from the horizontal deflection

Horizontal synchronization

The video input signal (positive video) is connected to pin 25. The horizontal synchronization has
two control loops that generate a sandcastle pulse. Using the oscillator sawtooth facilitates
accurate timing of the burst key pulse. Therefore, the phase of this sawtooth must have a fixed
relationship to the sync pulse, which is achieved by use of the second control loop.

The TDA8305A's horizontal synchronization circuit differs from that of the TDA4505 in that:

 The horizontal oscil|ator's retrace occurs during the horizontal retrace and not during the scan
period. This means that with weak input signals no interference will be visible on the screen.
It also prevents video crosstalk from disturbing the picture phase
 The reference signal for the horizontal phase detector is nearer to being symmetrical and is
independent of the supply voltage and temperature. As a consequence the frequency shift of
the horizontal oscillator during noise is reduced
 The current ratio of the phase detector for strong and weak signals is increased to obtain better
behaviour during both VCR-playback and weak signal reception. The switching level is independent
of supply voltage and temperature.

Horizontal phase detector
The circuit has the following operating conditions.

(a) Strong input signal, synchronized or non-synchronized.
(The strong/weak signal condition is obtained from the AGC circuit; the in-sync/out«of—sync
from the coincidence detector). In this condition the time constant is optimum for VCR-playback
i.e. fast time constant during the vertical retrace (to be able to correct VCR head-errors) and
such, that during scan, fluctuations of the sync are corrected. The phase detector is not gated.

lb) Weak signal — synchronized
In this condition the time constant is increased compared to condition (a). Also the phase
detector is gated when the oscillator is synchronized. This ensures a stable display which is not
disturbed by noise in the video signal.

(c) Weak signal - non-synchronized.

in this condition the time constant during scan and vertical retrace are the same as during scan
in condition (3).

Vertical synchronization
The TDA8305A's vertical circuit differs from that ofthe TDA4505 in that it has:
 Improved interlacing ~ the timing of the internal pulses is now close to a 50/50 ratio. This timing
is independent of supply voltage and temperature
 The temperature drift of the vertical amplitude has been reduced
 Reduction of noise in the vertical output signal so that modulation of the line distance will no
longer be visible on large screen sets.
 When out-of-sync is detected by the horizontal circuit the divider is switched to 625 lines.
This results in a stable amplitude when no input signal is available. In the TDA4505 the divider
remains in the wide window during this condition which means interference may affect stability.

Vertical sync pulse

The vertical sync pulse integrator will not be disturbed when the vertical sync pulses have a width
of 10 ,us with a separation of 22 ps. These types of vertical sync pulses are sometimes generated by
video tapes with anti-copy guard.

Vertical divider system

The TDA8305A embodies a synchronized divider system for generating the vertical sawtooth at pin 2.
The divider system has an internal frequency doubling circuit, which allows the horizontal oscillator to
operate at its normal line frequency. One line period equals 2 clock pulses.

Use of the divider system avoids the requirement for vertical frequency adjustment. The divider has
a discriminator window for automatic switching from 60 Hz to 50 Hz mode. When the trigger pulse
comes before line 576 the 60 Hz mode is selected, otherwise the 50 Hz mode is selected.

The divider system operates with 2 different divider reset windows for maximum interference}
disturbance protection.

The windows are activated via an up/down counter.

The counter increases its counter value by 1 each time the separated vertical sync pulse is within the
search window. When not within the search window this value is decreased by 1.

The operating modes of the divider system are as follows:

Mode A
Large search window (divider ratio between 488 and 722)

This mode is valid for the following conditions:

 Divider is looking for a new transmitter

 Divider ratio found - not within the narrow window limits

 Up/down counter value of the divider system operating in the narrow window mode drops below
count 10

Mode 8
Narrow window (divider ratio between 522 to 528, 60 Hz; or 622 to 628, 50 Hz)

The divider system switches over to this mode when the up/down counter has reached its maximum
value of 15 approved vertical sync pulses. when the divider operates in this mode and a vertical sync
pulse is missing within the window, the divider is reset at the end of the window and the counter value
is decreased by 1. At a counter value below 10 the divider system switches over to the large window

The divider system also generates an anti-topflutter pulse which inhibits the phase 1 detector during
the vertical sync pulse. The pulse width is dependent on the divider mode. In Mode A the start is
generated by reset of the divider
in Mode B the anti-topflutter pulse starts at the beginning of the first equalizing pulse.

The anti-topflutter pulse ends at count 10 for the 50 Hz mode and count 12 for the 60 Hz mode.

The vertical blanking pulse is also generated via the divider system. The start is by reset of the divider
while the blanking pulse width is 34 (17 lines) for the 60 Hz mode and at count 42 (21 lines) for
the 50 Hz mode.

The vertical blanking pulse at the sandcastle output (pin 27) is generated by adding the anti-topflutter
pulse to the blanking pulse. Thus the vertical blanking pulse starts at the beginning of the first
equalizing pulse when the divider operates in Mode B. The length of the vertical blanking in this
condition is 21 lines in the 60 Hz mode and 25 lines in the 50 Hz mode.

Application when external video signals require synchronization

The input to the sync separator is externally available via pin 25. For normal application the video
output signal at pin 17 is AC coupled to this input as shown in Fig.11. It is possible to interrupt this
connection and drive the sync separator from other sources such as:

 A teletext decoder in serial mode
 An external video signal via a peritelevision connector

When a teletext decoder is applied the IF amplifier and synchronization circuit are operating in the
same phase which means that various connections between the two sections li.e. AGC gating) can
remain active. When external signals are applied to the sync separator the connections between the
two parts must be interrupted. This can be achieved by connecting pin 22 to ground, which results
in the following conditions:

AGC detector is not gated

 AFC circuit is active

 Mute circuit not active — sound channel remains switched on

 Phase detector 1 has an optimal time constant for external video sources and is not gated.

X-ray protection

By forcing pin 1 below 1 V the horizontal output changes to a high resistance. The protection can be

released by switching off the mains.

The AFC control voltage is obtained by multiplying the IF output signal (which is also used to
drive the synchronous demodulator) with a reference carrier. This reference carrier is obtained
from the demodulator tuned circuit via a 90 degree phase shift network. The IF output signal
has an asymmetrical frequency spectrum with respect to the carrier frequency. To avoid
problems due to this asymmetrical signal the AFC circuit is followed by a sample-and-hold
circuit which samples during the sync level. As a result the AFC output voltage contains no
video information. The specified control steepness is without using an external load resistor.
The control steepness decreases when the AFC output is loaded with two resistors between
the voltage supply and ground.

At very weak input signals the drive signal for the AFC circuit will have a high noise content.
This noise input has an asymmetrical frequency spectrum which will cause an offset of the
AFC output voltage. To avoid problems due to this effect a notch filter can be built in to the
demodulator tuned circuit. The characteristics given for weak input signals are measured without
a notch circuit, with a SAW filter connected in front of the IC (input signal such that the input
signal of the IC is 150 uV (RMS value).

The minimum value is obtained with a 1.8 k9. series resistor connected between pin 17 and
pin 25. The slicing level can be varied by changing the value of this resistor (a higher resistance
results in a larger value of the minimum sync pulse amplitude). The slicing level is independent
of the video information.

Frequency control is obtained by supplying a correction current to the oscillator RC-network.
This is achieved via a resistor connected between the phase 1 detector output and the oscillator
network. The oscillator can be adjusted to the correct frequency by:
 short-circuit the sync separator bias network (pin 25) to the voltage supply.

To avoid the necessity of a VCR switch, the time constant of the phase detector at strong input
signals is sufficiently short to obtain a stable picture during VCFl playback. During the vertical
retrace period the time constant is even shorter so that VCR head errors are compensated for at
the beginning of the scan. During weak signal conditions (information derived from the AGC
circuit) the time constant is increased to obtain a good noise immunity.

This figure is valid for an external load impedance of 82 kfl connected between pin 28 and

the shift adjustment potentiometer.

The horizontal flyback input and the sandcastle output have been combined on pin 27. The
flyback pulse is clamped to a level of 4.5 V. The minimum current to drive the second control
loop is 0.1 mA.

The functions in—sync/out-of-sync and transmitter identification have been combined on pin 22.
The capacitor is charged during the sync pulse and discharged during the time difference
between gating and sync pulse.

The vertical scan is synchronized by means of a divider system, therefore no adjustment is
required for the ramp generator. The divider detects whether the incoming signal has a vertical
frequency of 50 or 60 Hz and corrects the vertical amplitude.

To avoid screenburn due to a collapse of the vertical deflection, a continuous blanking level is
inserted into the sandcastle pulse when the feedback voltage of the vertical deflection is not
within the specified limits.

PHILIPS TDA3565 PAL decoder:

The TDA3565 PAL decoder contains all the functions required for PAL signal decoding and colour matrixing and is contained within an 18-pin package. The oscillator, a.c.c. detector and burst phase detector each have single-pin outputs and the coupling capacitor for the luminance input at pin 8 doubles as a storage capacitor for the black level clamping circuit. Black level clamping of the three colour channels is performed using feedback proportional to the red channel black level. This feedback (variable with the brightness control) controls the input level of the luminance amplifier and therefore the clamping levels of all three colour signal outputs.

PHILIPS TDA3653B TDA3653C Vertical deflection and guard circuit  (90˚)

The TDA3653B/C is a vertical deflection output circuit for drive of various deflection systems with currents up to 1.5 A peak-to-peak.
• Driver
• Output stage
• Thermal protection and output stage protection
• Flyback generator
• Voltage stabilizer
• Guard circuit

Output stage and protection circuit
Pin 5 is the output pin. The supply for the output stage is fed to pin 6 and the output stage ground is connected to pin 4.
The output transistors of the class-B output stage can each deliver 0.75 A maximum.
The maximum voltage for pin 5 and 6 is 60 V.
The output power transistors are protected such that their operation remains within the SOAR area. This is achieved by
the co-operation of the thermal protection circuit, the current-voltage detector, the short-circuit protection and the special
measures in the internal circuit layout.
Driver and switching circuit Pin 1 is the input for the driver of the output stage. The signal at pin 1 is also applied via external resistors to pin 3 which is the input of a switching circuit. When the flyback starts, this switching circuit rapidly turns off the lower output stage and so limits the turn-off dissipation. It also allows a quick start of the flyback generator.
External connection of pin 1 to pin 3 allows for applications in which the pins are driven separately.
Flyback generator
During scan the capacitor connected between pins 6 and 8 is charged to a level which is dependent on the value of the resistor at pin 8
When the flyback starts and the voltage at the output pin (pin 5) exceeds the supply voltage, the flyback generator is activated.
The supply voltage is then connected in series, via pin 8, with the voltage across the capacitor during the flyback period.
This implies that during scan the supply voltage can be reduced to the required scan voltage plus saturation voltage of the output transistors.
The amplitude of the flyback voltage can be chosen by changing the value of the external resistor at pin 8.
It should be noted that the application is chosen such that the lowest voltage at pin 8 is > 2.5 V, during normal operation.
Guard circuit
When there is no deflection current and the flyback generator is not activated, the voltage at pin 8 reduces to less than 1.8 V. The guard circuit will then produce a DC voltage at pin 7, which can be used to blank the picture tube and thus prevent screen damage.
Voltage stabilizer
The internal voltage stabilizer provides a stabilized supply of 6 V to drive the output stage, which prevents the drive current of the output stage being affected by supply voltage variations.

PHONOLA (PHILIPS) 37KV1232  CHASSIS GR1AX  Switched-mode self oscillating supply voltage circuit:POWER SUPPLY (PHILIPS SOPS - Self Oscillating Power Supply)

A switched-mode self-oscillating supply voltage circuit for converting an input voltage into an output d.c. voltage which is substantially independent of variations of the input voltage and/or a load connected to the output voltage. The circuit comprises a first controllable switch connected in series with a transformer winding and a second controllable switch for turning-off the first switch. The conduction period of the first switch is controlled by means of a control voltage present on a control electrode of the second switch. The circuit can be switched-over to a stand-up state in which the energy supplied to the load is reduced to zero. A starting network is connected between the input voltage and the second switch so that the current therein flows through the second switch during the period of time this switch conducts and does not flow to the control electode of the first switch in the stand-by state.

1. A switched-mode self-oscillating supply voltage circuit for converting an input voltage into an output d.c. voltage which is substantially independent of variations of the input voltage and/or of a load connected to the terminals of the output voltage, comprising a transformer having a primary and a feedback winding, a first controllable switch connected in series with the primary winding, the series arrangement thus formed being coupled between terminals for the input voltage, a second controllable switch coupled via a turn-off capacitor to the control electrode of the first switch to turn it off, means coupling the feedback winding to said control electrode, a transformer winding being coupled via a rectifier to an output capacitor having terminals which supply the output voltage, an output voltage-dependent control voltage being present on a control electrode of the second switch for controlling the conduction period of the first switch, the circuit being switchable between an operating state and a stand-by state in which relative to the operating state the supply energy supplied to the load is considerably reduced, a starting network connected to a terminal for the input voltage, means for adjusting the control voltage in the stand-by state to a value at which the first controllable switch is cut-off, a connection which carries current during the conduction period for the second controllable switch being provided between the starting network and said second switch, and means providing a connection between the starting network and the control electrode of the first switch, which connection does not carry current in the stand-by state.

2. A supply voltage circuit as claimed in claim 1, further comprising a resistor included between the connection of the starting network to the second switch and a turn-off capacitor present in the connection to the control electrode of the first switch.

3. A supply voltage circuit as claimed in claim 2, characterized in that the second controllable switch comprises a thyristor having a main current path included in the control electrode connection of the first controllable switch, said thyristor having a first control gate electrode for adjusting the turn-off instant of the first switch and a second control electrode to which the starting network and the resistor are connected.

4. A supply voltage circuit as claimed in claim 1, characterized in that a resistor is included in the connection to the control electrode of the second controllable switch so that a current flows through said resistor in the stand-by state of a value sufficient to cut-off the first controllable switch.

The invention relates to a switched-mode self-oscillating supply voltage circuit for converting an input voltage into an output d.c. voltage which is substantially independent of variations of the input voltage and/or of a load connected to the terminals of the output voltage. This circuit comprises a transformer having a primary and a feedback winding and a first controllable switch arranged in series with the primary winding. The series arrangement thus formed is coupled between the terminals of the input voltage. A second controllable switch which is coupled via a turnoff capacitor to the control electrode of the first switch to turn it off. The feedback winding is coupled to this control electrode and the primary winding is coupled via a rectifier to an output capacitor the terminals of which are the terminals for the output voltage. An output voltage-dependent control voltage is present on a control electrode of the second switch for controlling the conduction period of the first switch. The circuit is switchable between an operating state and a stand-by state in which relative to the operating state the energy supplied to the load is considerably reduced, and the circuit further comprises a starting network connected to a terminal for the input voltage.
Such a supply voltage circuit is disclosed in German Patent Application No. 2,651,196. With this prior art circuit supply energy can be applied in the operating state to the different portions of a television receiver. In the stand-by state the majority of the output voltages of the circuit are so low that the receiver is substantially in the switched-off condition. In the prior art circuit the starting network is formed by a resistor connected to the unstabilized input voltage and through which on turn-on of the circuit a current flows via the feedback winding to the control electrode of the first controllable switch, which is a switching transistor, and brings it to and maintains it in the conductive state, as a result of which the circuit can start.
In the stand-by state the transistor is non-conducting in a large part of the period of the generated oscillation so that little energy is stored in the transformer. However, the starting resistor is connected via a diode to the second controllable switch, which is a thyristor. As the sum of the voltages across these elements is higher than the base-emitter threshold voltage of the transistor, the diode and the thyristor cannot simultaneously carry current. This implies that current flows through the starting resistor to the base of the transistor via the feedback winding after a capacitor connected to the feedback winding has been charged.
The invention has for its object to provide an improved circuit of the same type in which in the stand-by state the supply energy applied to the load is reduced to zero. The prior art circuit cannot be improved in this respect without the use of mechanical switches, for example relays. According to the invention, the switched-mode self-oscillating supply voltage circuit does not comprise such relays and is characterized in that it further comprises means for adjusting the control voltage in the stand-by state to a value at which the first controllable switch is cut-off. A connection which carries current during the conduction period of the second controllable switch is provided between the starting network and said second switch while a connection present between the starting network and the control electrode of the first switch does not carry current in the stand-by state.
The invention is based on the recognition that the prior art supply voltage circuit cannot oscillate, so that the energy supplied by it is zero, if the control voltage obtains a value as referred to, while the starting network is connected in such a manner that in the stand-by state no current can flow through it to the control electrode of the first controllable switch.
It should be noted that in the said German Patent Application the starting network is in the form of a resistor which is connected to an unstabilized input d.c. voltage. It is, however, known, for example, from German Patent Specification No. 2,417,628 to employ for this purpose a rectifier network connected to an a.c. voltage from which the said input d.c. voltage is derived by rectification.

The invention will now be further described by way of example with reference to the accompanying drawing, which shows a basic circuit diagram of a switched-mode self-oscillating supply voltage circuit.

The self-oscillating supply circuit shown in the FIGURE comprises a npn-switching transistor Tr1 having its collector connected to the primary winding L1 of a transformer T, while the emitter is connected to ground via a small resistor R1, for example 1.5 Ohm. Resistor R1 is decoupled for the high frequencies by means of a 150 nF capacitor C1. One end of winding L1 is connected to a conductor which carries an unstabilized input d.c. voltage V B of, for example, 300 V. Voltage V B has a negative rail connected to ground and is derived from the electric power supply by rectification. One end of a feedback winding L2 is connected to the base of transistor Tr1 via the parallel arrangement of a small inductance L3 and a damping resistor R2. A terminal of a 47 μF capacitor C2 is connected to the junction of the elements L2, L3 and R2. The series arrangement of a diode D1 and a 2.2 Ohm-limiting resistor R3 is arranged between the other terminal of capacitor C2 and the other end of winding L2 and the series arrangement of a resistor R4 of 12 Ohm and a diode D2 is arranged between the same end of winding L2 and the emitter of transistor Tr1. A 150 nF capacitor C3 is connected in parallel with diode D2. The anode of diode D1 is connected to that end of winding L2 which is not connected to capacitor C2, while the anode of diode D2 is connected to the emitter of transistor Tr1. In the FIGURE the winding sense of windings L1 and L2 is indicated by means of dots.
The junction of capacitor C2 and resistor R3 is connected to a 100 Ohm resistor R5 and to the emitter of a pnp-transistor Tr2. The base of transistor Tr2 is connected to the other terminal of resistor R5 and to the collector of an npn-transistor Tr3, whose emitter is connected to ground. The base of Tr3 is connected to the collector of transistor Tr2. Transistors Tr2 and Tr3 form an artificial thyristor, i.e. a controllable diode whose anode is the emitter of transistor Tr2 while the cathode is the emitter of transistor Tr3. The base of transistor Tr2 is the anode gate and the base of transistor Tr3 is the cathode gate of the thyristor formed. Between the last-mentioned base and the emitter of transistor Tr1 there is arranged the series network of a 2.2 kOhm resistor R6 with the parallel arrangement of a 2.2 kOhm resistor R7 and a 100 μF capacitor C4. The series arrangement of a diode D11 and a 220 Ohm limiting resistor R19 is arranged between the junction of components R6, R7 and C4 and the junction of components C2, L2, R2 and L3. The cathode of diode D11 is connected to capacitor C2.
Because of the feedback the described circuit oscillates independently as soon as the steady state is achieved. It will be described hereinafter how this state is obtained. During the time transistor Tr1 conducts the current flowing through the resistor R1 increases linearly. The resistor R4 then partly determines the base current of transistor Tr1. Capacitor C4 and resistor R7 form a voltage source the voltage of which is subtracted from the voltage drop across resistor R1. As soon as the voltage on the base of transistor Tr3 is equal to approximately 0.7 V this transistor becomes conductive, as a result of which the thyristor formed by transistors Tr2 and Tr3 becomes rapidly conductive and remains so. Across capacitor C2 there is a negative voltage by means of which transistor Tr1 is turned off. The inverse base current thereof flows through thyristor Tr2, Tr3. This causes charge to be withdrawn from capacitor C2, while the charge carriers stored in transistor Tr1 are removed with the aid of inductance L3. As soon as the collector current of transistor Tr1 has been turned off, the voltage across winding L2 reverses its polarity, which current recharges the capacitor. Now the voltage at the junction of components C2, R3 and R5 is negative, causing thyristor Tr2, Tr3 to extinguish.
Secondary windings L4, L5 and L6 are provided on the core of transformer T with the indicated winding senses. When transistor Tr1 is turned off, a current which recharges a smoothing capacitor C5, C6 or C7 via a rectifier D3, D4 or D5 flows through each of these windings. The voltages across these capacitors are the output voltages of the supply circuit for loads connectable thereto. These loads, which are not shown in the FIGURE, are, for example, portions of a television receiver.
In parallel with winding L1 there is the series network of a 2.2 nF tuning capacitor C8 and a 100 Ohm limiting resistor R8. The anode of a diode D6 is connected to the junction of components R8 and C8, while the cathode is connected to the other terminal of resistor R8. Winding L1 and capacitor C8 form a resonant circuit across which an oscillation is produced after windings L4, L5 and L6 have become currentless. At a later instant the current through circuit L1, C8 reverses its direction. As a result thereof a current is generated in winding L2 which flows via diode D2 and resistor R4 to the base of transistor Tr1 and makes this transistor conductive and maintains it in this state. The dissipation in resistor R8 is reduced by means of diode D6. A clamping network formed by the parallel arrangement of a 22 kOhm resistor R9 and a 120 nF capacitor C9 is arranged in series with a diode D7. This whole assembly is in parallel with winding L1 and cuts-off parasitic oscillations which would be produced during the period of time in which transistor Tr1 is non-conductive. The output voltages of the supply circuit are kept substantially constant in spite of variations of voltage V B and/or the loads, thanks to a control of the turning-on instant of thyrisistor Tr2, Tr3. For this purpose the emitter of a light-sensitive transistor Tr4 is connected to the base of transistor Tr3. The collector of transistor Tr4 is connected via a resistor R10 to the conductor which carries the voltage V B and to a Zener diode Z1 which has a positive voltage of approximately 7.5 V, while the base is unconnected. The other end of diode Z1 is connected to ground. A light-emitting diode D8, whose cathode is connected to the collector of an npn-transistor Tr5, is optically coupled to transistor Tr4. By means of a potentiometer R11 the base of transistor Tr5 can be adjusted to a d.c. voltage which is derived from the voltage V 0 of approximately 130 V across capacitor C6. The anode of diode D8 is connected to a d.c. voltage V 1 of approximately 13 V. A resistor R12 is also connected to voltage V 1 , the other end of the resistor being connected to the emitter of transistor Tr5, to the cathode of a Zener diode Z2 which has a voltage of approximately 7.5 V and to a smoothing capacitor C10. The other ends of diode Z2 and capacitor C10 are connected to ground. Voltage V1 can be generated by means of a transformer connected to the electric AC supply and a rectifier, which are not shown for the sake of simplicity, more specifically for a remote control to which constantly supply energy is always applied, even when the majority of the components of the receiver in what is referred to as the stand-by state are not supplied with supply energy.
A portion of voltage V 0 is compared with the voltage of diode Z2 by means of transistor Tr5. The measured difference determines the collector current of transistor Tr5 and consequently the emitter current of transistor Tr4. This emitter current produces across resistor R6 a voltage drop whose polarity is the opposite of the polarity of the voltage source formed by resistor R7 and capacitor C4. Under the influence of this voltage drop the turn-on instant of thyristor Tr2, Tr3 is controlled as a function of voltage V 0 . If, for example, voltage V 0 tends to decrease owing to an increasing load thereon and/or in response to a decrease in voltage V B , then the collector current of transistor Tr5 decreases and consequently also the said voltage drop. Thyristor Tr2, Tr3 is turned on at a later instant than would otherwise be the case, causing transistor Tr1 to be cut-off at a later instant. The final value of the collector current of this transistor is consequently higher. Consequently, the ratio of the time interval in which transistor Tr1 is conductive to the entire period, commonly referred to as the duty cycle, increases, while the frequency decreases.
The circuit is protected from overvoltage. This is ensured by a thyristor which is formed by a pnp-transistor Tr6 and an npn-transistor Tr7. The anode of a diode D9 is connected to the junction of components R3 and C2 and the cathode to the base of transistor Tr6 and to the collector of transistor Tr7. The base of transistor Tr7, which base is connected to the collector of transistor Tr6, is connected via a zener diode Z3 to a voltage which, by means of a potentiometer R13 is adjusted to a value derived from the voltage across capacitor C7. The emitter of transistor Tr6 also is connected to the voltage of capacitor C7, more specifically via a resistor R14 and a diode D10. If this voltage increases to above a predetermined value then thyristor Tr6, Tr7 becomes conductive. Since the emitter of transistor Tr7 is connected to ground, the voltage at its collector becomes very low, as a result of which diode D9 becomes conductive, which keeps transistor Tr1 in the non-conducting state. This situation is maintained as long as thyristor Tr6, Tr7 continues to conduct. This conduction time is predominantly determined by the values of capacitor C7, resistor R14 and a resistor R15 connected between the base and the emitter of transistor Tr6. A thyristor is advantageously used here to render it possible to switch off a large current even with a low level signal and to obtain the required hysteresis.
The circuit comprises a 1 MOhm starting resistor R16, one end of which is connected to the base of transistor Tr2 and the other end to the conductor which carries the voltage V B . Upon turn-on of the circuit current flows through resistors R16 and R5 and through capacitor C2, which has as yet no charge, to the base of transistor Tr1. The voltage drop thus produced across resistor R5 keeps transistor Tr2, and consequently also transistor Tr3, in the non-conductive state, while transistor Tr1 is made conductive and is maintained so by this current. Current also flows through winding L2. In this manner the circuit can start as energy is built up in transformer T.
The supply circuit can be brought into the stand-by state by making an npn-transistor Tr8, which is non-conductive in the operating state, conductive. The emitter of transistor Tr8 is connected to ground while the collector is connected to the collector of transistor Tr5 via a 1.8 kOhm resistor R17. A resistor R18 has one end connected to the base of transistor Tr8 and the other end, either in the operating state to ground, or in the stand-by state to a positive voltage of, for example, 5 V. Transistor Tr8 conducts in response to this voltage. An additional, large current flows through diode D8 and consequently also through transistor Tr4, resulting in thyristor Tr2, Tr3 being made conductive and transistor Tr1 being made non-conductive and maintained so. So to all appearances a large control current is obtained causing the duty cycle to be reduced to zero. A condition for a correct operation is that the emitter current of transistor Tr4 be sufficiently large in all circumstances, which implies that the voltage drop produced across resistor R6 by this current is always higher than the sum of the voltage across voltage source R7, C4, of the base-emitter threshold voltage of transistor Tr3 in the conductive state thereof, and of the voltage at the emitter of transistor Tr1. So the said voltage drop must be higher than the sum of the first two voltages, which corresponds to the worst dimensioning case in which the stand-by state is initiated while transistor Tr1 is in the non-conductive state.
If thyristor Tr2, Tr3 conducts, either in the operating state or in the stand-by state, current flows through resistor R16 via the collector emitter path of transistor Tr3 to ground. This current is too small to have any appreciable influence on the behaviour of the circuit. When thyristor Tr2, Tr3 does not conduct, the voltage on the left hand terminal of capacitor C2 is equal to approximately 1 V, while the voltage across the capacitor is approximately -4 V. So transistor Tr1 remains in the non-conductive state and a premature turn-on thereof cannot occur. If in the operating state transistor Tr1 conducts while thyristor Tr2, Tr3 is cut-off, then the current flows through resistor R16 in the same manner as it flows during the start to the base of transistor Tr1, but has relatively little influence as the base current caused by the energy stored in winding L2 is many times larger. If both transistor Tr1 and thyristor Tr2, Tr3 are non-conductive, then the current through resistor R16 flows through components R5, C2, L2, R4, C3 and R1. In this stand-by state capacitor C2 has indeed substantially no negative charge any longer but, in spite thereof, transistor Tr1 cannot become conductive since no current flows to its base. It will furthermore be noted that the circuit is protected in the event that thyristor Tr2, Tr3 has an interruption. Namely, in such a case the circuit cannot start.
In the foregoing a circuit is described which may be considered to be a switched-mode supply voltage circuit of the parallel ("flyback") type. It will be obvious that the invention may alternatively be used in supply voltage circuits of a different type, for example converters of the type commonly referred to as up-converters. It will also be obvious that transistor Tr1 may be replaced by an equivalent switch, for example a gate-turn-off switch.